Impedance tuning

ABSTRACT

The disclosure features wireless power transfer systems that include a power transmitting apparatus configured to wirelessly transmit power, a power receiving apparatus connected to an electrical load and configured to receive power from the power transmitting apparatus, and a controller connected to the power transmitting apparatus and configured to receive information about a phase difference between output voltage and current waveforms in a power source of the power transmitting apparatus, and to adjust a frequency of the transmitted power based on the measured phase difference.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of and claims priority to U.S.application Ser. No. 15/837,904, filed on Dec. 11, 2017, which is acontinuation of U.S. application Ser. No. 14/459,870, filed on Aug. 14,2014, which claims priority to U.S. Provisional Patent Application No.61/927,452, filed on Jan. 14, 2014, to U.S. Provisional PatentApplication No. 61/865,910, filed on Aug. 14, 2013, and to U.S.Provisional Patent Application No. 62/024,993, filed on Jul. 15, 2014,the entire contents of each of which are incorporated herein byreference.

TECHNICAL FIELD

This disclosure relates to wireless power transfer.

BACKGROUND

Energy or power may be transferred wirelessly using a variety of knownradiative, or far-field, and non-radiative, or near-field, techniques.For example, radiative wireless information transfer usinglow-directionality antennas, such as those used in radio and cellularcommunications systems and home computer networks, may be consideredwireless energy transfer. However, this type of radiative transfer isvery inefficient because only a tiny portion of the supplied or radiatedpower, namely, that portion in the direction of, and overlapping with,the receiver is picked up. The vast majority of the power is radiatedaway in all the other directions and lost in free space. Suchinefficient power transfer may be acceptable for data transmission, butis not practical for transferring useful amounts of electrical energyfor the purpose of doing work, such as for powering or chargingelectrical devices.

One way to improve the transfer efficiency of some radiative energytransfer schemes is to use directional antennas to confine andpreferentially direct the radiated energy towards a receiver. However,these directed radiation schemes may require an uninterruptibleline-of-sight and potentially complicated tracking and steeringmechanisms in the case of mobile transmitters and/or receivers. Inaddition, such schemes may pose hazards to objects or people that crossor intersect the beam when modest to high amounts of power are beingtransmitted. A known non-radiative, or near-field, wireless energytransfer scheme, often referred to as either induction or traditionalinduction, does not (intentionally) radiate power, but uses anoscillating current passing through a primary coil, to generate anoscillating magnetic near-field that induces currents in a near-byreceiving or secondary coil. Traditional induction schemes havedemonstrated the transmission of modest to large amounts of power,however only over very short distances, and with very small offsettolerances between the primary power supply unit and the secondaryreceiver unit. Electric transformers and proximity chargers are examplesof devices that utilize this known short range, near-field energytransfer scheme.

A need exists for a wireless power transfer scheme that is capable oftransferring useful amounts of electrical power over mid-range distancesor alignment offsets. Such a wireless power transfer scheme shouldenable useful energy transfer over greater distances and alignmentoffsets than those realized with traditional induction schemes, butwithout the limitations and risks inherent in radiative transmissionschemes.

SUMMARY

In general, in a first aspect, the disclosure features wireless powertransfer systems that include a power transmitting apparatus configuredto wirelessly transmit power, a power receiving apparatus connected toan electrical load and configured to receive power from the powertransmitting apparatus, and a controller connected to the powertransmitting apparatus and configured to receive information about aphase difference between output voltage and current waveforms in a powersource of the power transmitting apparatus, and adjust a frequency ofthe transmitted power based on the measured phase difference.

Embodiments of the systems can include any one or more of the followingfeatures.

The power receiving apparatus can be mounted on an electric vehicle. Theload can include one or more batteries of an electric vehicle. The loadcan include an electrical circuit or electrical system of a vehicle.

The controller can be configured to adjust the frequency to minimize thephase difference. The controller can be configured to adjust a busvoltage of the power source based on a target output power of the powertransmitting apparatus. The controller can be configured to adjust aphase control of the power source based on the target output power.

The controller can be configured to iteratively adjust the frequency anddetermine, after each iteration, whether the adjustment to the frequencyincreases or decreases the phase difference. The controller can beconfigured to adjust the frequency by a magnitude of at most 5% of anominal operating frequency of the wireless power transfer system.

Embodiments of the systems can also include any of the other featuresdisclosed herein, including features disclosed in connection withdifferent embodiments, in any combination as appropriate.

In another aspect, the disclosure features methods for wireless powertransfer that include using a power transmitting apparatus to wirelesslytransfer power at a selected frequency to a power receiving apparatusconnected to an electrical load to deliver power to the load, receivinginformation about a phase difference between output voltage and currentwaveforms generated by a power source in the power transmittingapparatus, and adjusting the frequency based on the phase difference.

Embodiments of the methods can include any one or more of the followingfeatures.

The power receiving apparatus can be mounted on an electric vehicle. Theload can include one or more batteries of an electric vehicle. The loadcan include an electrical circuit or electrical system of a vehicle.

The methods can include adjusting the frequency to determine a minimumvalue of the phase difference. The methods can include adjusting a busvoltage of a power source of the power transmitting apparatus based on atarget output power of the power transmitting apparatus. The methods caninclude adjusting a phase control value of the power source based on thetarget output power.

The methods can include iteratively adjusting the frequency anddetermining, after each iteration, whether the adjustment to thefrequency increases or decreases the phase difference. The methods caninclude adjusting the frequency by a magnitude of at most 5% of anominal frequency of the power transferred by the wireless powertransmitting apparatus.

Embodiments of the methods can also include any of the other steps orfeatures disclosed herein, including steps and features disclosed inconnection with different embodiments, in any combination asappropriate.

In a further aspect, the disclosure features detectors for use in awireless power transfer system, the detectors including a first inputterminal configured to receive a first electrical signal, a second inputterminal configured to receive a second electrical signal, a first logicunit connected to the first and second input terminals and configured toproduce a first output waveform based on the first and second electricalsignals, and a second logic unit connected to the first logic unit andconfigured to produce a second output waveform based on the first outputwaveform, where the second output waveform includes a pulse having awidth that corresponds to a temporal offset between the first and secondelectrical signals.

Embodiments of the detectors can include any one or more of thefollowing features.

The first electrical signal can correspond to a waveform representing anelectrical current in an amplifier of the system. The electrical currentcan corresponds to an output current of the amplifier.

The second electrical signal can correspond to a waveform representing avoltage in an amplifier of the system. The voltage can correspond to avoltage of a load coupled to the system.

The second output waveform can include a pulse having a width thatcorresponds to a temporal offset between the electrical current and thevoltage waveforms in the amplifier of the system. The pulse can have asquare wave profile.

The first logic unit can include an AND gate. The second logic unit caninclude an XOR gate.

The second logic unit can include a first terminal connected to thefirst logic unit and a second terminal, where the second logic unit isconfigured to receive the first output waveform at the first terminal,and the second electrical signal at the second terminal.

The detectors can include a measurement unit featuring a first terminalconnected to the second logic unit, where the measurement unit isconfigured to generate an output value that corresponds to the temporaloffset between the first and second electrical signals. The measurementunit can include a second terminal, and the measurement unit can beconfigured to receive a clock signal featuring a plurality of pulsesseparated by a constant temporal interval at the second terminal. Theoutput value can correspond to the temporal offset in a multiple of theconstant temporal interval.

The measurement unit can be configured to receive the second outputwaveform at the first terminal of the measurement unit, and themeasurement unit can be configured to increment a counter of clocksignal pulses when the second output waveform is positively-valued.

Embodiments of the detectors can also include any of the other featuresdisclosed herein, including features disclosed in connection with otherembodiments, in any combination, as appropriate.

In another aspect, the disclosure features methods of determining atemporal offset value between current and voltage waveforms in awireless power transfer system, the method including: performing a firstlogical operation on the current and voltage waveforms in the wirelesspower transfer system to generate a first output waveform, where thefirst logical operation corresponds to an AND operation; performing asecond logical operation on the first output waveform and the voltagewaveform in the wireless transfer system to generate a second outputwaveform, where the second logical operation corresponds to an XORoperation; and determining the temporal offset value based on the secondoutput waveform.

Embodiments of the methods can include any one or more of the followingfeatures.

The second output waveform can include a square waveform having a widththat corresponds to the temporal offset value. The methods can includemeasuring the width of the square waveform to determine the temporaloffset value. Measuring the width of the square waveform can includecounting a plurality of signal pulses during an interval thatcorresponds to the width of the square waveform, and outputting thecounted number of signal pulses, where the counted number of signalpulses corresponds to the temporal offset value.

Embodiments of the methods can also include any of the other steps orfeatures disclosed herein, including steps and features disclosed inconnection with other embodiments, in any combination, as appropriate.

In a further aspect, the disclosure features methods for assessing anoperating condition of a wireless power source, the methods includingdetermining whether the wireless power source is operating in acapacitive mode, and reducing the output power of the power source ifthe source is operating in a capacitive mode.

Embodiments of the methods can include any one or more of the followingfeatures.

The methods can include determining whether the wireless power source isoperating in a capacitive mode based on a temporal offset value betweencurrent and voltage waveforms in the power source. The current waveformcan correspond to an output current of an amplifier of the power source.The voltage waveform can correspond to a voltage of a load coupled tothe wireless power source.

The methods can include: (a) determining the temporal offset value; (b)comparing the measured temporal offset value to a first threshold value;and (c) determining that the power source is operating in a capacitivemode if the temporal offset value is less than the first thresholdvalue. The methods can include: repeating steps (a)-(c); determining acount of a number of consecutive determinations that the power source isoperating in a capacitive mode; comparing the count of the number ofconsecutive determinations that the power source is operating in acapacitive mode to a second threshold value; and reducing an outputpower of the power source if the count exceeds a second threshold value.

Determining the temporal offset value can include: performing a firstlogical operation on the current and voltage waveforms to generate afirst output waveform, where the first logical operation corresponds toan AND operation; performing a second logical operation on the firstoutput waveform and the voltage waveform to generate a second outputwaveform, where the second logical operation corresponds to an XORoperation; and determining the temporal offset value based on the secondoutput waveform.

The second output waveform can include a square waveform having a widththat corresponds to the temporal offset value, and the methods caninclude measuring the width of the square waveform to determine thetemporal offset value. Measuring the width of the square waveform caninclude counting a plurality of signal pulses during an interval thatcorresponds to the width of the square waveform. The methods can includeselecting the first threshold value based on a load coupled to thewireless power source.

The methods can include determining whether the wireless power source isoperating in a reactive mode, and reducing the output power of the powersource if the source is operating in a reactive mode. The methods caninclude determining whether the wireless power source is operating in areactive mode based on a temporal offset value between current andvoltage waveforms in the power source. The methods can includedetermining whether the wireless power source is operating in a reactivemode based on a magnitude of an output current of an amplifier of thepower source when a voltage generated by the amplifier changes polarity.The methods can include determining whether the wireless power source isoperating in a reactive mode based on a bus voltage in the amplifier.

The methods can include: determining the temporal offset value;determining a magnitude of the output current when the voltage generatedby the amplifier changes polarity; determining a bus voltage of theamplifier; comparing the temporal offset value to a first thresholdvalue; comparing the magnitude of the output current to a secondthreshold value; comparing the bus voltage to a third threshold value;and determining that the power source is operating in a reactive mode ifthe temporal offset value, the magnitude of the output current when thevoltage generated by the amplifier changes polarity, and the bus voltageof the amplifier exceed the first, second, and third threshold values,respectively.

Determining the temporal offset value can include: performing a firstlogical operation on the current and voltage waveforms to generate afirst output waveform, where the first logical operation corresponds toan AND operation; performing a second logical operation on the firstoutput waveform and the voltage waveform to generate a second outputwaveform, where the second logical operation corresponds to an XORoperation; and determining the temporal offset value based on the secondoutput waveform. The second output waveform can include a squarewaveform having a width that corresponds to the temporal offset value,and the methods can include measuring the width of the square waveformto determine the temporal offset value.

Embodiments of the methods can also include any of the other steps orfeatures disclosed herein, including steps and features disclosed inconnection with other embodiments, in any combination, as appropriate.

In another aspect, the disclosure features methods of transferring powerbetween a source resonator and a receiver resonator, the methodsincluding: setting an impedance and an output power level for the sourceresonator; determining whether the source resonator is operating in acapacitive mode; reducing the output power level if the source resonatoris operating in a capacitive mode; determining an efficiency of powertransfer between the source resonator and the receiver resonator;comparing the efficiency of power transfer to a threshold efficiencyvalue; and adjusting an impedance of the source resonator if theefficiency of power transfer is less than the threshold efficiencyvalue.

Embodiments of the methods can include any one or more of the followingfeatures.

The methods can include determining whether the source resonator isoperating in a reactive mode, and reducing the output power level if thesource resonator is operating in a reactive mode. Adjusting theimpedance of the source resonator can include electrically changing aninductance of a tunable inductor. Adjusting the impedance of the sourceresonator can include mechanically changing an inductance of a tunableinductor.

The methods can include transferring 1 kW or more (e.g., 3.3 kW or more)of power between the source resonator and the receiver resonator.

Determining whether the source resonator is operating in a capacitivemode can include: determining a temporal offset value between currentand voltage waveforms in the source resonator; comparing the measuredtemporal offset value to a first threshold value; and determining thatthe source resonator is operating in a capacitive mode if the temporaloffset value is less than the first threshold value. Determining thetemporal offset value can include: performing a first logical operationon the current and voltage waveforms to generate a first outputwaveform, where the first logical operation corresponds to an ANDoperation; performing a second logical operation on the first outputwaveform and the voltage waveform to generate a second output waveform,where the second logical operation corresponds to an XOR operation; anddetermining the temporal offset value based on the second outputwaveform.

The second output waveform can include a square waveform having a widththat corresponds to the temporal offset value, and the methods caninclude measuring the width of the square waveform to determine thetemporal offset value.

Determining whether the source resonator is operating in a reactive modecan include: determining a temporal offset value between current andvoltage waveforms in the source resonator; determining a magnitude of anoutput current of an amplifier in the source resonator when a voltagegenerated by the amplifier changes polarity; determining a bus voltageof the amplifier; comparing the temporal offset value to a firstthreshold value; comparing the magnitude of the output current to asecond threshold value; comparing the bus voltage to a third thresholdvalue; and determining that the power source is operating in a reactivemode if the temporal offset value, the magnitude of the output currentwhen the voltage generated by the amplifier changes polarity, and thebus voltage of the amplifier exceed the first, second, and thirdthreshold values, respectively.

Embodiments of the methods can also include any of the other steps orfeatures disclosed herein, including steps and features disclosed inconnection with other embodiments, in any combination, as appropriate.

In a further aspect, the disclosure features wireless power transfersystems that include any of the detectors disclosed herein, a sourceresonator featuring a coil having at least one loop of conductingmaterial, and an amplifier configured to generate an electrical current,where during operation, the systems are configured to transfer 2 kW ofpower or more (e.g., 4 kW of power or more) to a receiver resonator.

Embodiments of the systems can include any one or more of the followingfeatures.

During operation, the systems can be configured to transfer power to areceiver resonator positioned in a vehicle to charge a battery coupledto the receiver resonator.

Embodiments of the systems can also include any of the other featuresdisclosed herein, including features disclosed in connection with otherembodiments, in any combination, as appropriate.

In another aspect, the disclosure features wireless power transfersystems that include: a source resonator featuring a coil, where thesource resonator is configured to transfer power to a receivingresonator; a current generator coupled to the source resonator andconfigured to generate an electrical current in the coil; a detectorfeaturing a first input terminal configured to receive a first waveformcorresponding to the electrical current generated by the currentgenerator, a second input terminal configured to receive a secondwaveform corresponding to a voltage within the system, and at least onelogic unit, where the at least one logic unit is configured to generatean output waveform that includes a pulse having a width that correspondsto a temporal offset between the first and second waveforms; and anelectronic processor connected to the detector, where during operation,the electronic processor receives the output waveform and is configuredto determine whether the source resonator is operating in a capacitivemode based on the output waveform.

Embodiments of the systems can include any one or more of the featuresdisclosed herein, including features disclosed in connection with any ofthe embodiments disclosed herein, in any combination, as appropriate.

In the embodiments disclosed herein, a magnetic resonator can include acombination of inductors and capacitors. Additional circuit elementssuch as capacitors, inductors, resistors, switches, and the like, may beinserted between a magnetic resonator and a power source, and/or betweena magnetic resonator and a power load. In this disclosure, theconducting coil of the resonator may be referred to as the inductorand/or the inductive load. The inductive load may also refer to theinductor when it is wirelessly coupled (through a mutual inductance) toother system or extraneous objects. In this disclosure, circuit elementsother than the inductive load may be referred to as being part of animpedance matching network (IMN). In this disclosure, all, some, or noneof the elements that are referred to as being part of an impedancematching network may be part of the magnetic resonator. Which elementsare part of the resonator and which are separate from the resonator willdepend on the specific magnetic resonator and wireless energy transfersystem design.

In the wireless energy transfer systems described herein, power can beexchanged wirelessly between at least two resonators. Resonators cansupply, receive, hold, transfer, and distribute energy. Sources ofwireless power can be referred to as sources or supplies, and receiversof wireless power can be referred to as devices, receivers and/or powerloads. A resonator can be a source, a device, or both simultaneously,and/or may vary from one function to another in a controlled manner.Resonators configured to hold or distribute energy that do not havewired connections to a power supply or power drain can be calledrepeaters.

The resonators of the wireless energy transfer systems disclosed hereinare able to transfer power over distances that are large compared to thesize of the resonators. That is, if the resonator size is characterizedby the radius of the smallest sphere that could enclose the resonatorstructure, the wireless energy transfer systems disclosed herein cantransfer power over distances greater than the characteristic size ofthe resonator. The systems are able to exchange energy betweenresonators where the resonators have different characteristic sizes andwhere the inductive elements of the resonators have different sizes,different shapes, and/or are formed of different materials.

The wireless energy transfer systems disclosed herein can include morethan two resonators that can each be coupled to a power source, a powerload, both, or neither. Wirelessly supplied energy can be used to powerelectric or electronic equipment, recharge batteries, and/or chargeenergy storage units. Multiple devices can be charged or poweredsimultaneously, or power delivery to multiple devices can be serializedsuch that one or more devices receive power for a period of time afterwhich power delivery is switched to other devices. In some embodiments,multiple devices can share power from one or more sourcessimultaneously, or in a time multiplexed manner, or in a frequencymultiplexed manner, or in a spatially multiplexed manner, or in anorientation multiplexed manner, or in any combination of time and/orfrequency and/or spatial and orientation multiplexing. Multiple devicescan share power with one another, with at least one device beingreconfigured continuously, intermittently, periodically, occasionally,or temporarily, to operate as a wireless power source.

In some embodiments, systems adapted for wireless power transfer caninclude a tunable resonant amplifier circuit provided for driving aninductive load and having a varying impedance. The circuit can include aswitching amplifier with a variable duty cycle, an inductive load, aconnection between the inductive load and the switching amplifier withat least one tunable component, and a feedback loop for adjusting the atleast one tunable component and the duty cycle of the amplifier. Thefeedback loop can adjust the duty cycle of the amplifier and the atleast one tunable component to maintain substantially zero voltageswitching and zero current switching at the output of the amplifierunder different load conditions of the inductive load. The at least onetunable component can include a tunable capacitor and/or tunableinductor. The tunable capacitor and/or inductor can be in series or inparallel with the inductive load. The connection between the inductiveload and the switching amplifier can include more than one tunablecomponent. The switching amplifier can use a variable switchingfrequency. A bus voltage of the switching amplifier can be variable andused to control an amount of power delivered to the inductive load.

The feedback loop can include an impedance measuring facility. Thefeedback loop can include a processor configured to monitor an impedanceat an output of the switching amplifier and to compute an adjustment tothe variable duty cycle of the switching amplifier such that zerovoltage switching is substantially maintained. The processor may beconfigured to compute a second adjustment to at least one tunablecomponent such that zero current switching is substantially maintained.The inductive load may include a high-Q magnetic resonator. The circuitmay be used as a source in a wireless power transmission system.

Unless otherwise defined, all technical and scientific terms used hereinhave the same meaning as commonly understood by one of ordinary skill inthe art to which this disclosure belongs. Although methods and materialssimilar or equivalent to those described herein can be used in thepractice or testing of the subject matter herein, suitable methods andmaterials are described below. All publications, patent applications,patents, and other references mentioned herein are incorporated byreference in their entirety. In case of conflict, the presentspecification, including definitions, will control. In addition, thematerials, methods, and examples are illustrative only and not intendedto be limiting.

The details of one or more embodiments are set forth in the accompanyingdrawings and the description below. Other features and advantages willbe apparent from the description, drawings, and claims.

DESCRIPTION OF DRAWINGS

FIG. 1 is a schematic diagram of a wireless power transfer system.

FIG. 2A is a schematic diagram of a wireless power transfer system thatdoes not include an impedance matching network.

FIG. 2B is a schematic diagram of a wireless power transfer system thatincludes an impedance matching network.

FIG. 2C is a schematic diagram of a wireless power transfer system thatincludes a tunable power generator and an impedance matching network.

FIG. 3 is a schematic diagram of a power source that includes ahalf-bridge switching power amplifier.

FIG. 4 is a schematic diagram of a power source that includes afull-bridge switching amplifier.

FIG. 5 is a schematic diagram of a class D power amplifier.

FIG. 6 is a plot showing current and voltage waveforms and a measurementof an offset between the waveforms.

FIG. 7 is a flow chart showing a series of steps for detectingcapacitive mode operation in a source resonator.

FIG. 8 is a plot showing current and voltage waveforms and measurementof the output current at the voltage switching time.

FIG. 9 is a flow chart showing a series of steps for detecting reactivemode operation in a source resonator.

FIG. 10 is a flow chart showing a series of steps for performingimpedance tuning of a source resonator in a wireless power transfersystem.

FIG. 11 is a schematic diagram of a mode detector.

FIG. 12 is a schematic diagram of a power amplifier that includes a modedetector.

FIG. 13 is a schematic diagram of a portion of a wireless power transfersystem.

FIGS. 14A and 14B are schematic diagrams of a tunable inductor.

FIG. 14C is a schematic plot showing the inductance of the coil of FIG.14B.

FIGS. 15A and 15B are schematic diagrams showing an example of magneticmaterial in a tunable inductor.

FIG. 16A is a schematic diagram showing another view of the wirelesspower transfer system of FIG. 13.

FIG. 16B is a schematic diagram of a portion of the wireless powertransfer system shown in FIG. 16A.

FIGS. 17A-17D are images demonstrating impedance tuning of the wirelesspower transfer system of FIG. 13.

FIG. 18 is a schematic diagram of another embodiment of a tunableinductor.

FIGS. 19A and 19B are schematic diagrams of a further embodiment of atunable inductor.

FIG. 20 is an image of another embodiment of a tunable inductor.

FIG. 21 is a schematic diagram of a portion of a power transfer systemfeaturing a tunable element for adjusting the impedance of a resonatorof the system.

FIG. 22 is a schematic diagram of two wireless power transfer systemswith different displacements between source and receiver resonators.

FIG. 23 is a schematic diagram of an impedance matching networktopology.

FIGS. 24A-F are plots showing figure of merit (FOM) values in thek-V_(load) domain calculated for different tuning methods for theimpedance matching network topology of FIG. 23.

FIGS. 25A-E are plots showing optimal values of the tuning parameter inthe k-V_(load) domain calculated for different tuning methods for theimpedance matching network topology of FIG. 23.

FIGS. 26A-F are plots showing values of the bus voltage (V_(bus)) in thek-V_(load) domain calculated for different tuning methods for theimpedance matching network topology of FIG. 23.

FIGS. 27A-F are plots showing values of the input phase (φ) in thek-V_(load) domain calculated for different tuning methods for theimpedance matching network topology of FIG. 23.

FIGS. 28A-F are plots showing values of combined coil-to-coiltransmission and impedance matching network efficiency in the k-V_(load)domain calculated for different tuning methods for the impedancematching network topology of FIG. 23.

FIGS. 29A-F are plots showing values of power dissipated in the sourcein the k-V_(load) domain calculated for different tuning methods for theimpedance matching network topology of FIG. 23.

FIGS. 30A-F are plots showing values of power dissipated in the devicein the k-V_(load) domain calculated for different tuning methods for theimpedance matching network topology of FIG. 23.

FIG. 31 is a schematic diagram of a wireless power transfer system foruse with an electric vehicle.

FIG. 32 is a schematic diagram of a wireless power transmitter and awireless power receiver.

FIG. 33 is a schematic diagram of another wireless power transfersystem.

FIG. 34 is a flow chart showing a series of steps for implementingfrequency tuning in wireless power transfer systems.

FIG. 35 is a graph showing measured voltage, current, and phasedifference waveforms in a power source of a wireless power transmissionapparatus.

FIG. 36 is a flow chart showing a series of steps for regulating theoutput power of a wireless power transfer system at a pre-determinedpower level.

FIG. 37 is a flow chart showing a series of steps for implementingfrequency optimization in wireless power transfer systems.

FIG. 38 is a schematic diagram showing example impedance matchingnetworks in a wireless power transfer system.

Like reference symbols in the various drawings indicate like elements.

DETAILED DESCRIPTION Introduction—Wireless Power Transfer Systems

This disclosure relates to wireless power transfer using coupledelectromagnetic resonators. Important considerations for resonator-basedpower transfer include resonator efficiency and resonator coupling.Factors affecting wireless power transfer including, e.g., coupled modetheory (CMT), coupling coefficients and factors, quality factors (alsoreferred to as Q-factors), and impedance matching are discussed, forexample, in U.S. Patent Application Publication Nos. 2010/0237709,2010/0181843, and 2012/0119569, the entire contents of each of which areincorporated herein by reference.

For purposes of this disclosure, a resonator may be defined as aresonant structure that can store energy in at least two differentforms, where the stored energy oscillates between the two forms. Theresonant structure has a specific oscillation mode with a resonant(modal) frequency, f, and a resonant (modal) field. The angular resonantfrequency, ω, may be defined as ω=2πf, the resonant period, T, may bedefined as T=1/f=2π/ω), and the resonant wavelength, λ, may be definedas λ=c/f, where c is the speed of the associated field waves (light, forelectromagnetic resonators). In the absence of loss mechanisms, couplingmechanisms or external energy supplying or draining mechanisms, thetotal amount of energy stored by the resonator, W, would stay fixed, butthe form of the energy would oscillate between the two forms supportedby the resonator, where one form would be maximum when the other isminimum and vice versa.

For example, a resonator can be constructed such that the two forms ofstored energy are magnetic energy and electric energy. Further, theresonator can be constructed such that the electric energy stored by theelectric field is primarily confined within the structure while themagnetic energy stored by the magnetic field is primarily in the regionsurrounding the resonator. In other words, the total electric andmagnetic energies would be equal, but their localization is different.Using such structures, energy exchange between at least two structurescan be mediated by the resonant magnetic near-field of the at least tworesonators. These types of resonators may be referred to as magneticresonators.

An important parameter of resonators used in wireless power transmissionsystems is the Quality Factor, or Q-factor, or Q, of the resonator,which characterizes the energy decay and is inversely proportional toenergy losses of the resonator. It may be defined as Q=ω*W/P, where P isthe time-averaged power lost at steady state. That is, a resonator witha high-Q has relatively low intrinsic losses and can store energy for arelatively long time. Since the resonator loses energy at its intrinsicdecay rate, 2Γ, its Q, also referred to as its intrinsic Q, is given byQ=ω/2Γ. The quality factor also represents the number of oscillationperiods, T, it takes for the energy in the resonator to decay by afactor of e. Note that the quality factor or intrinsic quality factor orQ of the resonator is that due only to intrinsic loss mechanisms. The Qof a resonator connected to, or coupled to a power generator, g, orload, l, may be called the “loaded quality factor” or the “loaded Q”.The Q of a resonator in the presence of an extraneous object that is notintended to be part of the energy transfer system may be called the“perturbed quality factor” or the “perturbed Q”.

Resonators having substantially the same resonant frequency, coupledthrough any portion of their near-fields, may interact and exchangeenergy. By way of example, but not limitation, imagine a sourceresonator with Q_(s), and a device resonator with Q_(d). High-Q wirelessenergy transfer systems may utilize resonators that are high-Q. The Q ofeach resonator may be high. The geometric mean of the resonator Q's,√{square root over (Q_(s)Q_(d))}, may also or instead be high.

The coupling factor, k, is a number between 0≤k≤1, and it may beindependent (or nearly independent) of the resonant frequencies of thesource and device resonators, when those are placed at sub-wavelengthdistances. Rather, the coupling factor k may be determined mostly by therelative geometry and the distance between the source and deviceresonators, where the physical decay-law of the field mediating theircoupling is taken into account. The coupling coefficient used in CMT,κ=κ√{square root over (ω_(s)ω_(d))}/2, may be a strong function of theresonant frequencies, as well as other properties of the resonatorstructures.

In applications for wireless energy transfer utilizing the near-fieldsof the resonators, it is desirable to have the size of the resonator bemuch smaller than the resonant wavelength, so that power lost byradiation is minimized. In some embodiments, high-Q resonators aresub-wavelength structures. In some embodiments, high-Q resonatorstructures are designed to have resonant frequencies higher than 50 kHz.In certain embodiments, the resonant frequencies may be less than 1 GHz.For example, in certain applications such as car charging, the resonantfrequencies are between 50 KHz and 500 KHz. In other applications, suchas charging consumer electronics, the resonant frequencies are, forexample, between 1 MHz and 1 GHz.

The power radiated into the far-field by sub-wavelength resonators canbe further reduced in some embodiments by lowering the resonantfrequency of the resonators and the operating frequency of the system.In certain embodiments, the far field radiation can be reduced byarranging for the far fields of two or more resonators to interferedestructively in the far field.

In wireless power transfer systems, a resonator can be used as awireless power source, a wireless power capture device, a repeater or acombination thereof. In some embodiments, a resonator can alternatebetween transferring power, receiving power, and/or relaying power. Inwireless power transfer systems, one or more magnetic resonators may becoupled to a power source and be energized to produce an oscillatingmagnetic near-field. Other resonators that are within the oscillatingmagnetic near-fields can capture these fields and convert the power intoelectrical energy that may be used to power or charge a load therebyenabling wireless transfer of useful power.

The so-called “useful” power in a useful power exchange is the powerthat is delivered to a device to power or charge it at an acceptablerate. The transfer efficiency that corresponds to a useful powerexchange may be system or application-dependent. For example, high powervehicle charging applications that transfer kilowatts of power may needto be at least 80% efficient to supply useful amounts of power resultingin a useful energy exchange sufficient to recharge a vehicle batterywithout significantly heating up various components of the transfersystem. In some consumer electronics applications, a useful powerexchange can include any power transfer efficiencies greater than 10%,or any other amount acceptable to keep rechargeable batteries “toppedoff” and running for long periods of time. In implanted medical deviceapplications, a useful power exchange can be any exchange that does notharm the patient but that extends the life of a battery or wakes up asensor or monitor or stimulator. In such applications, 100 mW of poweror less may be useful. In distributed sensing applications, powertransfer of microwatts may be useful, and transfer efficiencies may bewell below 1%.

In the embodiments disclosed herein, resonators may be referred to assource resonators, device resonators, first resonators, secondresonators, repeater resonators, and the like. Embodiments can includethree or more resonators. For example, a single source resonator cantransfer power to multiple device resonators and/or multiple devices.Power can be transferred from a first device to a second, and then fromthe second device to the third, and so forth. Multiple sources cantransfer power to a single device or to multiple devices connected to asingle device resonator or to multiple devices connected to multipledevice resonators.

Resonators can serve alternately or simultaneously as sources, devices,and/or they may be used to relay power from a source in one location toa device in another location. Intermediate electromagnetic resonatorscan be used to extend the distance range of wireless power transfersystems and/or to generate areas of concentrated magnetic near-fields.Multiple resonators can be daisy-chained together, exchanging power overextended distances and with a wide range of sources and devices. Forexample, a source resonator can transfer power to a device resonator viaseveral repeater resonators. Energy from a source can be transferred toa first repeater resonator, the first repeater resonator can transferthe power to a second repeater resonator and the second to a third, andso on, until the final repeater resonator transfers its power to adevice resonator. In this respect, the range or distance of wirelesspower transfer may be extended and/or tailored by adding repeaterresonators. High power levels may be split between multiple sources,transferred to multiple devices, and/or recombined at a distantlocation.

FIG. 1 shows a schematic diagram of an example embodiment of a wirelesspower transfer system. The system includes at least one source resonator(R1) 104 (and optionally, another source resonator R6, 112) coupled to apower source 102 and optionally to a sensor and control unit 108. Powersource 102 can be a source of any type of power capable of beingconverted into electrical energy that can be used to drive the sourceresonator 104. The power source can be a battery, a solar panel, theelectrical mains, a wind or water turbine, an electromagnetic resonator,and/or a generator. The electrical power used to drive the magneticresonator is converted into oscillating magnetic fields by theresonator. The oscillating magnetic fields can be captured by otherresonators which can be device resonators (R2) 106, (R3) 116 that areoptionally coupled to a power drain 110.

The oscillating fields can be optionally coupled to repeater resonators(R4, R5) that are configured to extend or tailor the wireless powertransfer topology. Device resonators can capture the magnetic fields inthe vicinity of source resonator(s), repeater resonators and otherdevice resonators and convert them into electrical power that may beused by a power drain. The power drain 110 can be an electrical,electronic, mechanical or chemical device and the like configured toreceive electrical energy. Repeater resonators can capture magneticfields in the vicinity of source, device, and repeater resonator(s) andmay transfer the power on to other resonators.

Wireless power transfer systems can include a single source resonator104 coupled to a power source 102 and a single device resonator 106coupled to a power drain 110. In some embodiments, wireless powertransfer systems can include multiple source resonators coupled to oneor more power sources and can include multiple device resonators coupledto one or more power drains.

Power can be transferred directly between source resonator 104 anddevice resonator 106. Alternatively, power can be transferred from oneor more source resonators 104, 112 to one or more device resonators 106,116 via any number of intermediate resonators which may be deviceresonators, source resonators, and/or repeater resonators. Power can betransferred via a network or arrangement of resonators 114 that caninclude sub-networks 118, 120 arranged in any combination of topologiessuch as, for example, token ring, mesh, and ad hoc.

In some embodiments, the wireless power transfer system can include acentralized sensing and control system 108. Parameters of theresonators, power sources, power drains, network topologies, andoperating parameters can be monitored and adjusted using one or moreprocessors in control system 108 to meet specific operating criteria ofthe system. For example, one or more central control processors canadjust parameters of individual components of the system to optimizeglobal power transfer efficiency, and/or to optimize the amount of powertransferred.

In some embodiments, the wireless power transfer system can have adistributed sensing and control system in which sensing and control canbe incorporated into each resonator or group of resonators, powersources, and power drains. The distributed sensing and control systemcan be configured to adjust the parameters of the individual componentsin the group to maximize the power delivered and/or to maximize powertransfer efficiency in that group, for example.

In certain embodiments, components of the wireless power transfer systemcan have wireless or wired data communication interfaces and/or links toother components such as devices, sources, repeaters, power sources, andresonators. The components can transmit and/or receive data using thelinks and/or interfaces that can be used to enable distributed orcentralized sensing and control. Wireless communication channels can beseparate from wireless energy transfer channels, or the same channel canbe used to perform both functions. In some embodiments, resonators usedfor power exchange can also be used to exchange information. Forexample, information can be exchanged by modulating a component in asource or device circuit and sensing that change with port parameter orother monitoring devices. Resonators can signal each other by tuning,changing, varying, dithering, and the like, the resonator parameterssuch as the impedance of the resonators which may affect the reflectedimpedance of other resonators in the system. The systems and methodsdisclosed herein enable the simultaneous transmission of power andcommunication signals between resonators in wireless power transmissionsystems, and enable the transmission of power and communication signalsduring different time periods and/or at different frequencies using thesame magnetic fields that are used during wireless power transfer. Incertain embodiments, wireless communication between components of thesystems disclosed herein can occur via a separate wireless communicationchannel, examples of which include WiFi, Bluetooth, and infraredchannels.

Impedance Matching in Wireless Power Transfer Systems

A variety of factors influence the amount of power that can betransferred wirelessly and the efficiency with which power istransferred from one resonator to another. In particular, the efficiencyof power transfer between coupled high-Q magnetic resonators is impactedby how closely matched the resonators are in resonant frequency and howwell their impedances are matched to the power supplies and powerconsumers (i.e., power drains) in the system. Because a variety ofexternal factors including the relative positions of extraneous objectsand/or other resonators in the system, and the changing of thoserelative positions, can alter the resonant frequency and/or inputimpedance of a high-Q magnetic resonator, wireless power transfersystems can include tunable components for adjusting the impedance ofresonators in the system to maintain target levels of power transmissionin various environments or operating scenarios.

Impedance tuning in systems that provide for wireless power transfer canbe accomplished by adjusting inductive elements of the systems. Forpurposes of the present discussion, an inductive element can be any coilor loop structure (the “loop”) of a conducting material, with or withouta core made of magnetic material (gapped or ungapped), which may also becoupled inductively or in any other contactless way to other systems.The element is inductive because its impedance has positive reactance,X, and resistance, R.

As an example, consider an external circuit, such as a driving circuitor a driven load or a transmission line, to which an inductive elementis connected. The external circuit (e.g. a driving circuit) can deliverpower to the inductive element and the inductive element can deliverpower to the external circuit (e.g., a driven load). The efficiency andamount of power delivered between the inductive element and the externalcircuit at a desired frequency can depend on the impedance of theinductive element relative to the properties of the external circuit.Impedance-matching networks and external circuit control techniques canbe used to regulate the power delivery between the external circuit andthe inductive element, at a desired frequency, f.

In general, maximum power is delivered from a source to a receiver(e.g., a load) when the impedance of the source, Z_(o), is the complexconjugate of the impedance of the receiver, Z_(o)*. However, achievingmaximum efficiency of power transfer does not require conjugate matchingof the impedances of the source and receiver. Typically, for example,power sources such as switching amplifiers that are used for wirelesspower transfer have very low impedance.

Accordingly, in many applications, an impedance matching networkconnected to an inductive element is used to adjust the input impedanceso that the efficiency of the amplifier is high, and the amplifierdelivers a target amount of power to an external circuit (e.g., areceiver). That is, the impedance matching network adjusts Z_(o) so thatthe amplifier (e.g., an amplifier of class A, B, C, D, DE, E, F and thelike) efficiently transfers a target amount of power to a receiver(e.g., a receiving resonator). The amount of power delivered can becontrolled by adjusting the complex ratio of the impedance of thecombination of the impedance matching network and the inductive elementat the power transfer frequency.

In general, impedance matching networks connected to inductive elementscan form magnetic resonators. For some applications, such as wirelesspower transfer using strongly-coupled magnetic resonators, a high Q maybe desired for the resonators. Therefore, the inductive element may beconfigured to have low losses (e.g., high X/R). Since the matchingcircuit can include additional sources of loss inside the resonator, thecomponents of the matching circuit can also be chosen to have lowlosses. Furthermore, in high-power applications and/or due to the highresonator Q, large currents may be present during operation in parts ofthe resonator circuit and large voltages may be present across somecircuit elements within the resonator. Such currents and voltages canexceed the specified tolerances for particular circuit elements and maybe too high for particular components to withstand. In some cases, itmay be difficult to find or implement components, such as tunablecapacitors for example, with size, cost and performance (loss andcurrent/voltage-rating) specifications sufficient to realize high-Q andhigh-power resonator designs for certain applications. However, suitablyconfigured impedance matching networks can preserve the high Q formagnetic resonators, while also reducing the component requirements forlow loss and/or high current/voltage-rating.

For example, impedance matching circuit topologies can be implemented toreduce or even minimize the loss and current-rating requirements on someof the elements of the matching circuit. The topology of a circuitmatching a low-loss inductive element to a target impedance, Z_(o), canbe implemented so that some of its components lie outside the associatedhigh-Q resonator by being in series with the external circuit. Therequirements for low series loss or high current ratings for thesecomponents can therefore be reduced. Relieving the low series lossand/or high-current-rating requirements on a circuit element may beparticularly useful when the element is variable and/or has a largevoltage rating.

As another example, impedance matching circuit topologies can beimplemented to reduce or even minimize the voltage rating requirementson some of the elements of the matching circuit. The topology of acircuit matching a low-loss inductive element to a target impedance,Z_(o), may be chosen so that some of its components lie outside theassociated high-Q resonator by being in parallel with Z_(o). As aresult, the requirements for low parallel loss and/or highvoltage-rating for these components may be reduced. Relieving the lowparallel loss and/or high-voltage requirement on a circuit element maybe particularly useful when the element needs to be variable and/or tohave a large current-rating and/or low series loss.

The topology of the circuit matching a low-loss inductive element to aparticular target impedance, Z_(o), may be chosen so that the fieldpattern of the associated resonant mode and thus its high Q arepreserved upon coupling of the resonator to the external impedance.Otherwise, inefficient coupling to the desired resonant mode may occur(potentially due to coupling to other undesired resonant modes),resulting in an effective lowering of the resonator Q.

For applications where the low-loss inductive element or the externalcircuit may exhibit variations, dynamic adjustment of the matchingcircuit can be performed to match the inductive element to a targetimpedance Z_(o) at the desired frequency f Since there may typically betwo tuning objectives, matching or controlling both the real andimaginary part of the impedance, Z_(o), at the desired frequency, f,there may be two variable elements in the matching circuit. Forinductive elements, the matching circuit may need to include at leastone variable capacitive element.

In some embodiments, a low-loss inductive element may be matched bytopologies using two variable capacitors, or two networks of variablecapacitors. A variable capacitor may, for example, be a tunablebutterfly-type capacitor having, e.g., a center terminal for connectionto a ground or other lead of a power source or load, and at least oneother terminal across which a capacitance of the tunable butterfly-typecapacitor can be varied or tuned, or any other capacitor having auser-configurable, variable capacitance.

In certain embodiments, a low-loss inductive element may be matched bytopologies using one, or a network of, variable capacitor(s) and one, ora network of, variable inductor(s). In some embodiments, a low-lossinductive element may be matched by topologies using one, or a networkof, variable capacitor(s) and one, or a network of, variable mutualinductance(s), which transformer-couple the inductive element either toan external circuit or to other systems.

In some embodiments, it may be difficult to find or implement tunablelumped elements with size, cost, and performance specificationssufficient to realize high-Q, high-power, and potentially high-speed,tunable resonator designs. The topology of the circuit matching avariable inductive element to an external circuit may be designed sothat some of the variability is assigned to the external circuit byvarying the frequency, amplitude, phase, waveform, duty cycle, and thelike, of the drive signals applied to transistors, diodes, switches andthe like, in the external circuit.

In certain embodiments, variations in resistance, R, and inductance, L,of an inductive element at the resonant frequency may be only partiallycompensated or not compensated at all. Adequate system performance maythus be preserved by tolerances designed into other system components orspecifications. Partial adjustments, realized using fewer tunablecomponents or less capable tunable components, may be sufficient.

In some embodiments, impedance matching circuit architectures areimplemented that achieve the desired variability of the impedancematching circuit under high-power conditions, while minimizing thevoltage/current rating requirements on its tunable elements andachieving a finer (i.e. more precise, with higher resolution) overalltunability. The topology of the circuit matching a variable inductiveelement to a target impedance, Z_(o), can include appropriatecombinations and placements of fixed and variable elements, so that thevoltage/current requirements for the variable components can be reducedand the desired tuning range can be covered with finer tuningresolution. In this manner, the voltage/current performance requirementscan be reduced on components that are not variable.

In general, impedance matching network architectures can be implementedto achieve a variety of objectives, including:

(1) to increase or even maximize the power delivered to, and/or toreduce or even minimize impedance mismatches between, the sourcelow-loss inductive elements (and any other systems wirelessly coupled tothem) from the power source(s) (e.g., power driving generators);

(2) to increase or even maximize the power delivered from, and/or toreduce or even minimize impedance mismatches between, the devicelow-loss inductive elements (and any other systems wirelessly coupled tothem) and the power driven loads;

(3) to deliver a controlled amount of power to, or to achieve a certainimpedance relationship between, the source low-loss inductive elements(and any other systems wirelessly coupled to them) and the power drivinggenerators; and

(4) to deliver a controlled amount of power from, or to achieve acertain impedance relationship between, the device low-loss inductiveelements (and any other systems wirelessly coupled to them) to the powerdriven loads.

As explained above, a wireless power source can include at least oneresonator coil coupled to a power supply, which may be a switchingamplifier, such as a class-D amplifier or a class-E amplifier, or acombination thereof. In such an arrangement, the resonator coil iseffectively a power load to the power supply. In some embodiments, awireless power device can include at least one resonator coil coupled toa power load, which may be a switching rectifier, such as a class-Drectifier or a class-E rectifier, or a combination thereof. In such anarrangement, the resonator coil is effectively a power supply for thepower load, and the impedance of the load directly relates also to thework-drainage rate of the load from the resonator coil.

In general, in any of these configurations, the efficiency of powertransmission between a power supply and a power load is impacted by howclosely matched the input impedance of the power load is to an impedancethat the power source can efficiently drive. Designing and/or adjustingthe power supply and/or power load impedance to obtain maximum powertransmission efficiency is typically referred to as “impedancematching,” and corresponds to optimizing the ratio of useful-to-lostpower in a wireless power transfer system.

As explained above, impedance matching can be performed by addingnetworks or sets of elements such as capacitors, inductors,transformers, switches, resistors, and the like, to form impedancematching networks between a power supply and a power load. Impedancematching can then be achieved via mechanical adjustments and changes inelement positioning. For varying loads, impedance matching networks caninclude variable components that are dynamically adjusted to ensure thatthe impedance at the power supply terminals looking towards the load andthe characteristic impedance of the power supply remain substantiallycomplex conjugates of each other, even in dynamic environments andoperating scenarios.

More generally, a variety of different adjustments can be performed toachieve impedance matching between a power supply and a load. In someembodiments, for example, impedance matching can be accomplished bytuning one or more of the duty cycle, the phase, and the frequency ofthe driving signal of the power supply. In certain embodiments,impedance matching can be achieved by tuning a physical component withinthe power supply, such as a capacitor. Such a tuning mechanism may beadvantageous because it may allow impedance matching between a powersupply and a load without the use of a tunable impedance matchingnetwork, or with a simplified tunable impedance matching network, suchas one that has fewer tunable components, for example. Tuning the dutycycle and/or frequency and/or phase of the driving signal to a powersupply can yield a dynamic impedance matching system with an extendedtuning range or precision, with higher power, voltage and/or currentcapabilities, with faster electronic control, and/or with fewer externalcomponents.

In addition to providing for impedance matching between a power load andpower supply at the time a wireless power transfer system is designed orset up, impedance tuning permits dynamic adjustment of the system'simpedance characteristics to account for changes in impedance that occurduring operation of the system. For example, in some wireless powertransfer systems, parameters of resonators such as the inductance may beaffected by environmental conditions such as surrounding objects,temperature, orientation, and number and position of other resonators.Changes in operating parameters of the resonators may change certainsystem parameters, such as the efficiency with which power istransferred between resonators. As an example, high-conductivitymaterials located near a resonator may shift the resonant frequency of aresonator and detune it from other resonant objects. To allow the systemto respond to such environmental changes, a resonator feedback mechanismcan be implemented that corrects the resonant frequency of the resonatorby changing a reactive element (e.g., an inductive element or capacitiveelement).

Active tuning circuits that include tunable components and that monitorthe operating environment and operating parameters of certain systemcomponents can be integrated into certain wireless power transfersystems. Monitoring circuits can generate signals to actively compensatefor changes in parameters of components. For example, temperaturemeasurements can be used to calculate expected changes in, or toindicate previously measured values of, the capacitance of the systemallowing compensation by switching in other capacitors or tuningcapacitors to maintain the desired capacitance over a range oftemperatures. In certain embodiments, RF amplifier switching waveformscan be adjusted to compensate for component value or load changes in thesystem. In some embodiments, changes in parameter values for certaincomponents can be compensated with active cooling, heating, activeenvironment conditioning, and the like.

The concepts discussed above are illustrated schematically in FIGS.2A-2C. As described, the efficiency of power transmission between apower generator and a power load can be impacted by how closely matchedthe input impedance of the load is to the desired impedance of thegenerator (e.g., an impedance at which the generator delivers power withhigh, or even maximal, efficiency). In the example embodiment of awireless power transfer system shown in FIG. 2A, power can betransferred from power generator 602 to power load 604 at a maximumpossible efficiency when the input impedance of load 604 is equal to thedesired impedance of the power generator 602 (which can also be a poweramplifier). Impedance matching between power generator 602 and powerload 604 can be performed by inserting a tunable impedance matchingnetwork 606 (which can include one or more sub-networks and/or sets ofelements such as capacitors, resistors, inductors, transformers,switches and the like) between the power generator and the power load,as shown in FIG. 2B.

In some embodiments, as explained further above, mechanical adjustmentsand/or changes in element positioning can be used to achieve impedancematching. The impedance matching network 606 can include variablecomponents that are dynamically adjusted in this manner to ensure thatthe impedance at the generator terminals looking towards the power loadremain substantially equal to the desired impedance of the powergenerator, even in dynamic environments and operating scenarios. Dynamicimpedance matching can be accomplished by tuning the duty cycle and/orthe phase and/or the frequency of the driving signal of a tunable powergenerator 608, as shown in FIG. 2C, and/or by tuning a physicalcomponent within the tunable power generator. Where a tunable powergenerator 608 is implemented as shown in FIG. 2C, impedance matching canbe performed with or without a tunable impedance matching network 606between the power generator and power load. Where an impedance matchingnetwork is present, the impedance matching network can, in certainembodiments, be simplified relative to the tunable impedance matchingnetwork in FIG. 2B, e.g., by including fewer tunable components.Further, tuning the duty cycle and/or frequency and/or phase of thedriving signal to a power generator may yield a dynamic impedancematching system with an extended tuning range or precision, with higherpower, voltage and/or current capabilities, with faster electroniccontrol, and/or with fewer external components.

A number of other considerations can be important for impedance matchingin wireless power transfer systems. In such systems, low loss inductiveelement is typically the coil of a source resonator coupled to one ormore device resonators or other resonators, such as repeater resonators,for example. The impedance of the inductive element, R+jωL, can includethe reflected impedances of the other resonators on the coil of thesource resonator. Variations of R and L of the inductive element canoccur due to external perturbations in the vicinity of the sourceresonator and/or the other resonators, and/or thermal drift ofcomponents. Variations of R and L of the inductive element can alsooccur during normal use of the wireless power transmission system due torelative motion of the devices and other resonators with respect to thesource. The relative motion of these devices and other resonators withrespect to the source, or relative motion or position of other sources,can lead to varying coupling (and thus varying reflected impedances) ofthe devices to the source. Furthermore, variations of Rand L of theinductive element can also occur during normal use of the wireless powertransmission system due to changes within the other coupled resonators,such as changes in the power draw of their loads. The methods andsystems disclosed herein are capable of compensating for such variationsby implementing dynamic impedance matching of between the inductiveelement and the external circuit driving it.

Consider, for example, a source resonator that includes a low-losssource coil which is inductively coupled to the device coil of a deviceresonator driving a resistive load. In some embodiments, dynamicimpedance matching can be achieved at the source circuit. In certainembodiments, dynamic impedance matching can also be achieved at thedevice circuit. The effective resistance of the source inductive element(namely the resistance of the source coil R_(s) plus the reflectedimpedance from the device) depends on the mutual inductance of thecoils. Similarly, the effective resistance of the device inductiveelement also depends on the mutual inductance. Dynamic variation of themutual inductance between the coils due to motion results in a dynamicvariation of the effective impedances. Therefore, when both source anddevice are dynamically tuned, the variation of mutual inductance is seenfrom the source circuit side as a variation in the source inductiveelement resistance R. Note that in this type of variation, the resonantfrequencies of the resonators may not change substantially, since L maynot be changing. Thus, the methods and systems disclosed herein fordynamic impedance matching can be used for the source circuit of thewireless power transmission system.

Note that the possible wireless power transmission efficiency alsoincreases with U. To achieve an approximately constant level of powertransmitted to the device, the output power of the source may need todecrease as U increases. If dynamic impedance matching is implemented bytuning some of the amplifier parameters, the output power of theamplifier may vary accordingly. In some embodiments, the automaticvariation of the output power is preferred to be monotonicallydecreasing with R, so that it matches the constant device powerrequirement. In embodiments where the output power level is accomplishedby adjusting the DC driving voltage of the power generator, using animpedance matching set of tunable parameters which leads tomonotonically decreasing output power vs. R allows constant power to bemaintained at the power load in the device with only a moderateadjustment of the DC driving voltage. In embodiments where adjustment ofthe output power level occurs via adjustment of the duty cycle DC or thephase of a switching amplifier or a component within an impedancematching network, using an impedance matching set of tunable parametersthat leads to monotonically decreasing output power vs. R allowsconstant power to be maintained at the power load in the device withonly a moderate adjustment of duty cycle or phase.

FIG. 3 is a block diagram of a power source that includes a half-bridgeswitching power amplifier and associated measurement, tuning, andcontrol circuitry. FIG. 4 is a block diagram of a power source thatincludes a full-bridge switching amplifier and associated measurement,tuning, and control circuitry. The half bridge system topology depictedin FIG. 3 includes a processing unit 328 that executes one or morecontrol algorithms. Processing unit 328 can be a micro controller, anapplication specific circuit, a field programmable gate array, aprocessor, a digital signal processor, and the like. The processing unitcan be a single device or it can be implemented as a network of devices.The control algorithm(s) can run on any portion of the processing unit.The algorithm(s) can be customized for certain applications and caninclude a combination of analog and digital circuits and signals. Thealgorithm(s) can measure and adjust voltage signals and levels, currentsignals and levels, signal phases, digital count settings, and the like.

The system shown in FIG. 3 also includes an optional source/deviceand/or source/other resonator communication controller 332 coupled towireless communication circuitry 312. The optional source/device and/orsource/other resonator communication controller 332 can be part of thesame processing unit that executes the control algorithm(s), it can bepart of a circuit within a micro controller 302, it can be external tothe wireless power transmission modules, and it can be substantiallysimilar to communication controllers used in wire powered or batterypowered applications but adapted to include new and/or differentfunctionality to enhance or support wireless power transmission.

The system includes a pulse-width modulation (PWM) generator 306 coupledto at least two transistor gate drivers 334 and controlled by thecontrol algorithm implemented on processing unit 328. The two transistorgate drivers 334 are coupled directly or via gate drive transformers totwo power transistors 336 that drive the source resonator coil 344through impedance matching network components 342. The power transistors336 can be coupled and powered with an adjustable DC supply 304 and theadjustable DC supply 304 can be controlled by a variable bus voltage,Vbus. The Vbus controller 326 may be controlled by the control algorithmand may be part of, or integrated into, microcontroller 302 or otherintegrated circuits. Vbus controller 326 controls the voltage output ofadjustable DC supply 304, which in turn controls the power output of theamplifier and the power delivered to the resonator coil 344.

The system shown in FIG. 3 also includes optional sensing andmeasurement circuitry including signal filtering and buffering circuits318, 320 that can shape, modify, filter, process, and buffer signalsprior to their input to processors and/or converters such as analog todigital converters (ADC) 314, 316, for example. The processors andconverters such as ADCs 314, 316 can be integrated into microcontroller302 or can be implemented as separate circuits that can be coupled to aprocessing core 330. Based on measured signals, the control algorithmcan generate, limit, initiate, extinguish, control, adjust, and/ormodify the operation of any of PWM generator 306, communicationcontroller 332, Vbus control 326, source impedance matching controller338, filtering/buffering elements 318 and 320, converters 314 and 316,and resonator coil 344. The impedance matching networks 342 andresonator coils 344 can include electrically controllable, variable,and/or tunable components such as capacitors, switches, and/orinductors, and these components can have their component values oroperating points adjusted according to signals received from the sourceimpedance matching controller 338.

Components can be tuned to adjust the operation and characteristics ofthe resonator including the power delivered to and by the resonator, theresonant frequency of the resonator, the impedance of the resonator, theQ of the resonator, and any other coupled systems. The resonator can beany of a variety of different types or structures of resonator includinga capacitively loaded loop resonator, a planar resonator with a magneticmaterial, and combinations thereof.

The full bridge system topology shown in FIG. 4 can include a processingunit 328 that executes a master control algorithm. Processing unit 328can be a microcontroller, an application specific circuit, a fieldprogrammable gate array, a processor, a digital signal processor, andthe like. The system can include a source/device and/or source/otherresonator communication controller 332 coupled to wireless communicationcircuitry 312. The source/device and/or source/other resonatorcommunication controller 332 can be part of the same processing unitthat executes the master control algorithm, or part of a circuit withina micro controller 302, or external to the wireless power transmissionmodules, and may be substantially similar to communication controllersused in wire powered or battery powered applications but adapted toinclude new and/or different functionality to enhance or supportwireless power transmission.

The system in FIG. 4 includes PWM generator 410 with at least twooutputs coupled to at least four transistor gate drivers 334 that can becontrolled by signals generated from the master control algorithm. Thefour transistor gate drivers 334 can be coupled to four powertransistors 336 directly or via gate drive transformers that can drivesource resonator coil 344 through impedance matching networks 342. Thepower transistors 336 can be coupled and powered with an adjustable DCpower supply 304, and power supply 304 is controlled by Vbus controller326, which is in turn controlled by the master control algorithm. Vbuscontroller 326 adjusts the voltage output of the adjustable DC powersupply 304, which controls the power output of the amplifier and powerdelivered to the resonator coil 344.

The system of FIG. 4 can optionally include sensing and measurementcircuitry including signal filtering and buffering circuits 318, 320 anddifferential/single ended conversion circuitry 402, 404 that can shape,modify, filter, process, and buffer signals prior to being input toprocessors and/or converters such as analog to digital converters (ADC)314, 316. The processors and/or converters, such as ADCs 314 and 316,can be integrated into microcontroller 302, or can be implemented asseparate circuits that are coupled to processing core 330. Based onmeasured signals, the master control algorithm can generate, limit,initiate, extinguish, control, adjust, and/or modify the operation ofany of PWM generator 410, communication controller 332, Vbus controller326, source impedance matching controller 338, filtering/bufferingelements 318 and 320, differential/single ended conversion circuitry 402and 404, converters 314 and 316, and resonator coil 344.

Impedance matching networks 342 and resonator coils 344 can includeelectrically controllable, variable, and/or tunable components such ascapacitors, switches, and inductors, and these components can have theircomponent values or operating points adjusted according to signalsreceived from source impedance matching controller 338. Components canbe tuned to enable tuning of the operation and characteristics of theresonator including the power delivered to and by the resonator, theresonant frequency of the resonator, the impedance of the resonator, theQ of the resonator, and any other coupled systems. The resonator can beany of a variety of different types or structures of resonator,including a capacitively loaded loop resonator, a planar resonator witha magnetic material, and combinations thereof.

Referring to FIGS. 3 and 4, impedance matching networks 342 can includefixed value components such as capacitors, inductors, and networks ofsuch components. Portions of the impedance matching networks can includeinductors, capacitors, transformers, and series and parallelcombinations of such components. In some embodiments, portions of theimpedance matching networks can be empty (i.e., short-circuited).

The full bridge topology shown in FIG. 4 can allow operation at higheroutput power levels, relative to the half-bridge topology shown in FIG.3, using the same DC bus voltage. The half-bridge topology of FIG. 3 canprovide a single-ended drive signal, while the full bridge topology ofFIG. 4 can provide a differential drive signal to source resonator 308.In some wireless power transfer applications, the driven load may havean impedance that is very different from the desired impedance of theexternal driving circuit to which it is connected. Furthermore, thedriven load may not be a resonant network. Impedance matching networks342 in FIGS. 3 and 4 regulate the impedance at the input of the networkconsisting of the impedance matching network (IMN) circuit and the load.An IMN circuit can achieve this regulation by creating a resonance closeto the driving frequency. Since such an IMN circuit accomplishes some oreven all conditions needed to increase or even maximize the powertransmission efficiency from the generator to the load (resonance andimpedance matching—ZVS and ZCS for a switching amplifier), the IMNcircuit can be used between the driving circuit and the load.

Where the load is variable and/or the inductance of the resonators inthe system varies due to environmental factors such as variations in therelative positions of the resonators, the presence of other componentsthat perturb the resonators, and changes in physical conditions such astemperature that change the material properties of the resonators,impedance matching between the load and the external driving circuit,such as a linear or switching power amplifier, can be achieved by usingadjustable/tunable components in the IMN circuit that may be adjusted inresponse to the varying load and/or changing resonator properties. Toadjust both the real and imaginary parts of the impedance, two (or, moregenerally, more than one, such as two or more) tunable/variable elementsin the IMN circuit can be used.

In some embodiments, the load may be inductive (such as a resonatorcoil) with impedance R+jωL, so the two tunable elements in the IMNcircuit may be two tunable capacitance networks, or one tunablecapacitance network and one tunable inductance network, or one tunablecapacitance network and one tunable mutual inductance network.

Further, in some embodiments, impedance matching for a linear orswitching power amplifier can be achieved by using adjustable/tunablecomponents or parameters in the amplifier circuit that may be adjustedto match the desired impedance of the amplifier to the varying inputimpedance of the network consisting of the IMN circuit and the load(IMN+load). As shown in FIG. 2C, the IMN circuit can also be tunable. Tomatch both the real and imaginary parts of the impedance, a total of twotunable/variable elements or parameters in the amplifier and the IMNcircuit can be used. As explained above, the number of tunable/variableelements in the IMN circuit can be reduced or even completely eliminatedby incorporating such elements in the amplifier. For example, onetunable element in the power amplifier and one tunable element in theIMN circuit can be used for impedance matching. As another example, twotunable elements in the power amplifier (and none in the IMN circuit)can be used for impedance matching. Tunable elements or parameters inthe power amplifier can include the frequency, amplitude, phase,waveform, and duty cycle of the drive signals applied to transistors,switches, diodes, and other circuit elements.

FIG. 5 shows a simplified circuit diagram of a class D power amplifier802, impedance matching network 804, and an inductive load 806.Switching amplifier 804 includes a power source 810, switching elements808, and capacitors. The impedance matching network 804 includesinductors and capacitors, and the load 806 is represented as an inductorand a resistor. Amplifier 802 can correspond to either a half-bridge orfull bridge class-D amplifier operating at switching frequency f anddriving a low-loss inductive element R+jωL via IMN 804.

In some embodiments, L′—the inductance of the inductive element in IMN804—can be tunable. L′ can be tuned, for example, via a variable tappingpoint on the inductor or by connecting a tunable capacitor in series orin parallel to the inductor. In some embodiments, C_(o) can be tunable.For the half-bridge topology, C_(o) can be tuned by varying either oneor both capacitors as only the parallel sum of these capacitors mattersfor the amplifier operation. For the full bridge topology, C_(o) can betuned by varying one, two, three, or all capacitors as only theircombination (series sum of the two parallel sums associated with the twohalves of the bridge) matters for the amplifier operation.

In some embodiments, two of the components IMN 804 can be tunable. Forexample, both L′ and C₂ can be tuned. FIG. 6 shows values of L′ and C₂that can be used to achieve impedance matching as functions of thevarying R and L of the inductive element, and the associated variationof the output power (at given DC bus voltage) of the amplifier, forf=250 kHz, dc=40%, C_(o)=640 pF, and C₁=10 nF. Since IMN 804 alwaysadjusts to the fixed desired impedance of the amplifier, the outputpower is constant as the inductive element is varying.

Capacitive and Reactive Operational Modes

Impedance tuning can be particularly important in wireless powertransfer applications where the positions of the source and receiverresonators vary with respect to one another. One such application iswireless transfer of power for vehicle charging. Although the receiverresonator in a vehicle is positioned in approximately the same locationwith respect to a source resonator configured for power transfer to thevehicle, the relative positions of the source and receiver resonatorsstill vary each time they are aligned with respect to one another (e.g.,when the vehicle's operator parks the vehicle over the sourceresonator). These variations in relative position between the source andreceiver resonators can lead to an impedance mismatch between the sourceand receiver resonators.

To understand the origin of the impedance mismatch, consider that whenthe vehicle's wireless power transfer system is initially designed, thedesign assumes a particular separation and relative orientation betweenthe source and receiver resonators. The impedance of the source andreceiver are matched based on this assumption. During operation, if therelative orientations of the source and receiver resonators differ fromthe assumed orientation, the efficiency with which power is transferredbetween them can be reduced.

Other factors can also contribute to reduced power transfer efficiencybetween the source and receiver resonators. In vehicle chargingapplications, debris may be located in the general vicinity of thesource and receiver resonators, which can alter the impedance of theresonators. Further, variations in environmental conditions (e.g.,variations in temperature, humidity) can lead to changes in theproperties of materials from which the source and receiver resonatorsare formed, which in turn can lead to changes in the impedances of theresonators. Still further, variations in the load represented by thereceiver resonator can occur, e.g., as batteries in the vehicle areclose to capacity after a period of charging, such that power transferefficiency between the source and receiver resonators is reduced duringcharging, even when efficiency was higher the onset of charging.

The systems and methods for impedance matching disclosed herein areparticularly useful for achieving a target impedance within the sourceto yield efficient wireless power transfer between the source andreceiver resonators. By achieving higher efficiency, the amount of powerthat can be transferred in a particular period of time also increases.

Transferring large amounts of power efficiently is an importantconsideration for many applications, including vehicle chargingapplications. Vehicle batteries have high capacities relative to thebatteries found in most conventional electronic devices. Moreover,consumer preferences and use patterns typically demand that even whensubstantially depleted, vehicle batteries are recharged within arelatively brief period of time. The systems and methods for impedancematching disclosed herein allow for wireless power transfer in highpower applications such as vehicle charging by permitting even largeamounts of power to be transferred at high efficiency between source andreceiver resonators.

Another aspect that affects the efficiency and amount of wireless powertransfer between source and receiver resonators, particularly in highpower applications such as vehicle charging, is the mode in which thesource and/or receiver resonators are operating. The followingdiscussion will focus on the source resonator, but it should beappreciated that the aspects and features discussed are also applicableto the receiver resonator in a wireless power transfer system.

As the impedance of a resonator (e.g., a source resonator) is tuned,particularly in high power applications such as vehicle charging, e.g.,where the amount of power transferred is 1 kW or more (e.g., 2 kW ormore, 3.3 kW or more, 4.5 kW or more, 5.5 kW or more, 6.4 kW or more),impedance matching to achieve a particular target impedance is not theonly criterion that affects efficient transfer of large amounts of powerbetween the source and receiver resonators. The impedance mode can alsosignificantly affect power transfer between the source and receiverresonators.

In general, as the coupling between source and receiver resonatorsincreases, the real component of the complex impedance of the sourceresonator increases, while the imaginary component of the compleximpedance decreases. When the imaginary component is greater than zero,the source resonator operates in an “inductive” or “reactive” mode;conversely, when the imaginary component is less than zero, the sourceresonator operates in a “capacitive” mode.

For high power wireless power transfer applications, operating thesource resonator (e.g., an amplifier) in a capacitive mode can lead tocertain problems. In particular, prolonged operation of the sourceresonator in a capacitive mode can lead to excessive switching lossesand/or possible component failures, in particular, failure of inverterpower switches. Potential catastrophic damage to the source resonatorcan result from such component failures.

Conversely, operating the source resonator in a mode that is tooinductive or reactive can also lead to inefficient power transfer.Reactive mode operation can occur when the relative positions of sourceand receiver coils lead to large circulating currents in the source(e.g., amplifier) output circuitry. These large currents can lead toexcessive power dissipation in the amplifier, significantly reducing theamount of power that is transferred to the receiver resonator.

To ensure that the source resonator does not operate in a capacitivemode, the systems disclosed herein detect possible capacitive modeoperation during start-up of the source resonator (e.g., during start-upof the power amplifier). In general, a variety of different methods canbe used to detect capacitive mode operation. In some embodiments, todetect capacitive mode operation, the systems include a mode detectorthat measures a time difference between the output voltage and currentwaveforms of the power amplifier. If the voltage and current waveformsare too close together (e.g., if their phase offset is not largeenough), the source resonator is determined to be operating incapacitive mode.

FIG. 6 is a schematic diagram showing output voltage and currentwaveforms from a power amplifier. In the systems and methods disclosedherein, a mode detector is used to measure the time difference 1010 (or,equivalently, the phase difference) between the current and voltagewaveforms. Capacitive mode operation can then be detected based on themeasured time difference. In particular, in FIG. 6, if the timedifference 1010 between the current and voltage waveforms is too small,the power amplifier is determined to be operating in capacitive mode.

FIG. 7 is a flow chart 1100 showing a series of steps for detectingcapacitive mode operation in a source resonator such as a poweramplifier. In step 1102, a mode detector in the system is initialized toprepare for measurement of the voltage-current time difference. In someembodiments, for example, initialization can be performed by setting (orre-setting) the value of a counter to zero. The counter is then used todetermine the voltage-current time difference, e.g., the offset betweenthe voltage and current waveforms can be measured in terms of counterincrements, each of which represents a time interval.

Also in step 1102, the value of a capacitive mode counter is set tozero. The capacitive mode counter counts the number of times thatcapacitive mode operation of the source resonator is detected.

Next, in step 1104, the mode detector is used to determine the offsetbetween the voltage and current output waveforms in the source resonator(e.g., power amplifier). The offset can be measured in a variety ofways; examples of methods and components for measuring the offset willbe discussed in a later section. It should be noted, however, that theoffset can be measured in various forms. In some embodiments, asdiscussed above, the offset can be measured in increments of a counter,where each counter increment corresponds to a particular time period(e.g., an internal clock cycle of the source resonator). In certainembodiments, the offset can be measured in units of time, either bydirect measurement of the offset, or by multiplying the offset incounter increments by the period of time corresponding to a singlecounter increment. In some embodiments, the offset can be measured inangular units, e.g., as a phase angle difference between the current andvoltage waveforms.

In step 1106, the measured offset is compared to a threshold offsetvalue that indicates capacitive operation. The threshold value isexpressed in the same units (e.g., counter increments, units of time,phase angle difference) as the measured offset. If the offset value isgreater than the threshold offset value, then it is determined thatbecause the current and voltage waveforms are far enough offset from oneanother, the source resonator is not operating in a capacitive mode.Control passes to optional step 1108, where the output power of thesource resonator (e.g., the power that is transferred wirelessly to thereceiver resonator) can be increased. Next, the mode detector is resetat step 1110 (e.g., the offset value is reset to zero), and controlreturns to step 1104 for another measurement of the current-voltageoffset.

If the offset value is smaller than the threshold offset value in step1106, then it is determined that because the current and voltagewaveforms are offset from one another by only a relatively small amount,the source resonator is operating in a capacitive mode. Control passesto step 1112, where the output of the source resonator is maintained.Then, in step 1114, the value of the capacitive mode counter isincremented. As explained above, the capacitive mode counter counts thenumber of times that capacitive mode operation of the source resonatorhas been detected.

Next, in step 1116, the value of the capacitive mode counter is comparedto a consecutive capacitive mode threshold value. The consecutivecapacitive mode threshold value represents a limit on the number oftimes that capacitive mode operation of the source resonator can bedetected before operation of the source resonator is modified. If thecapacitive mode counter value exceeds the consecutive capacitive modethreshold value (e.g., the number of consecutive times that capacitiveoperation of the source resonator has been detected exceeds thethreshold value represented by the consecutive capacitive modethreshold), then the output power of the source resonator is reduced instep 1118, or operation of the source resonator is halted. If the outputpower is merely reduced without halting operation of the sourceresonator, control can optionally return to step 1102.

Alternatively, if the capacitive mode counter value is less than theconsecutive capacitive mode threshold value in step 1116, then controlpasses to step 1110 where the mode detector is reset, and then to step1104 for another measurement of the voltage-current offset.

In step 1116 of flow chart 1110, the source resonator is evaluated forcapacitive mode operation by comparing the voltage-current offset to thecapacitive mode threshold value. In some embodiments, the capacitivemode threshold value remains fixed for different operating conditions ofthe source and receiver resonators. More generally, however, thecapacitive mode threshold value can be varied depending upon one oroperating parameters or conditions of the wireless power transfersystem. For example, in certain embodiments, the capacitive modethreshold value can vary based on the load voltage (e.g., the voltagemaintained by the source resonator across a device connected to thereceiver resonator. Table 1 provides a list of example capacitive modethreshold values as a function of the load voltage.

TABLE 1 Load Voltage (V) Capacitive Mode Threshold Value (ns) <300 342.8300-340 314.3 340-370 285.7 370-400 257.1 >400 257.1

The wireless power transfer systems disclosed herein can detect reactivemode operation of a source resonator (e.g., power amplifier). Detectionof reactive mode operation can occur in systems that also detectcapacitive mode operation, or alternatively, in systems where detectionof capacitive mode operation does not occur. As discussed above, whenthe source resonator of a wireless power transfer system operates in amode that is too reactive (e.g., too inductive), excessive powerdissipation can occur in the source resonator, which significantlyreduces the efficiency of wireless power transfer from the sourceresonator to the receiver resonator.

FIG. 8 is a schematic diagram showing output voltage and currentwaveforms from a power amplifier. The same mode detector that measuresthe time difference (or phase difference) between the current andvoltage waveforms for purposes of detecting capacitive mode operationcan also be used to measure the source resonator (e.g., amplifier)output current at the voltage switching time, 1210. The current-voltageoffset, the output current at the voltage switching time 1210, and thesource resonator (e.g., amplifier) bus voltage can then be used todetect reactive mode operation of the source resonator.

FIG. 9 is a flow chart 1300 showing a series of steps for detectingreactive mode operation in a source resonator such as a power amplifier.In step 1302, the mode detector is initialized as described above inconnection with FIG. 7. Next, in step 1304, the mode detector is used tomeasure the offset (e.g., phase difference) between the output currentand voltage waveforms of the source resonator (e.g., power amplifier).In general, the offset is measured in step 1304 in the same manner as instep 1104 of FIG. 7.

In step 1306, the output current at the voltage switching time ismeasured by the mode detector, or by another sensor in the system.Referring to FIG. 8, the voltage switching time is the time at which thesign of the voltage switches. The mode detector or other sensorsmeasures the output current at the time when the voltage polaritychanges.

Returning to FIG. 9, in step 1308, bus voltage of the source resonator(e.g., power amplifier) is measured by the mode detector or by anothersensor or element in the system.

To determine whether the source resonator is operating in a reactivemode (e.g., a mode where the inductance is so high that significantpower dissipation occurs), the measured bus voltage is compared to athreshold bus voltage value in step 1310. If the measured bus voltage isless than the bus voltage threshold value, then it is determined thatthe source resonator is not operating in a reactive mode in step 1320,and the procedure terminates at step 1322. Alternatively, if themeasured bus voltage is greater than the bus voltage threshold value,then it is possible that the source resonator is operating in a reactivemode, and control passes to step 1312.

In step 1312, the measured current-voltage offset is compared to acapacitance mode detection threshold value. If the current-voltageoffset is less than the capacitance mode detection threshold value, thenit is determined that the source resonator is not operating in areactive mode in step 1320, and the procedure terminates at step 1322.However, if the current-voltage offset is greater than the capacitancemode detection threshold value, then it is possible that the sourceresonator is operating in a reactive mode, and control passes to step1314.

In step 1314, the measured output current at the voltage switching timeis compared to a threshold value for the output current at the voltageswitching time. If the measured output current is smaller than thethreshold current value, then it is determined that the source resonatoris not operating in a reactive mode in step 1320, and the procedureterminates at step 1322. However, if the measured output current at thevoltage switching time is larger than the threshold value in step 1314(which also implies that the measured bus voltage is larger than thethreshold bus voltage in step 1310, and the measured current-voltageoffset is larger than the capacitance mode detection threshold value instep 1312), then the source resonator (e.g., power amplifier) isdetermined to be operating in a reactive mode in step 1316.

In optional step 1318, the output power of the source resonator can bereduced so that the source resonator operates in a mode that is lessreactive. The procedure then terminates at step 1322.

Values for the threshold bus voltage, capacitance mode detectionthreshold, and threshold output current at the voltage switching timecan be obtained in various ways. In some embodiments, for example,threshold values can be hard-coded into the system's logic processor(s).In certain embodiments, some or all of the threshold values can bestored in a storage unit and retrieved when the system is powered on. Insome embodiments, some or all of the threshold values can be supplied byan operator of the system.

In general, each of the thresholds can have a variety of values based onfactors such as the tolerances and operating limits of the components ofthe system, the operating environment of the system, and the wirelesspower transfer application. Further, as discussed above, each of thethresholds can have multiple values according to certain operatingparameters such as the load voltage, output power, etc. As an example,in some embodiments, the source resonator is determined to be operatingin a reactive mode if the bus voltage exceeds a threshold value of 400V, if the current-voltage offset exceeds a capacitance mode detectionthreshold value of 857.1 ns, and the output current at the voltageswitching time exceeds a threshold value of 20 A.

Capacitive mode detection and/or reactive mode detection can beincorporated into control algorithms for a variety of wireless powertransfer systems, including the systems disclosed herein. In general,detection of capacitive and/or reactive mode operation occurs during orafter impedance tuning to simultaneously ensure that the impedancebetween the source and receiver resonators is matched as closely aspossible, while at the same time ensuring that operation in a mode thatis either too capacitive or too reactive is avoided. By performingimpedance tuning in conjunction with capacitive and/or reactive modedetection, power can be wirelessly transferred between the source andreceiver resonators at high efficiency, and at the same time, failure ofcomponents of the source resonator (e.g., inverter switches) can beavoided.

FIG. 10 is a flow chart 1400 showing a series of steps for performingimpedance tuning of a source resonator in a wireless power transfersystem. In the first step 1402, the initial frequency, impedance, andoutput power of the source resonator are set. Initial values for each ofthese operating parameters can be set according to stored information(e.g., stored values of the parameters) and/or according to user input,for example.

Next, in optional step 1404, the system determines whether the sourceresonator is operating in a capacitive mode. Some or all of the stepsdisclosed in connection with FIG. 7 can be used to make thisdetermination, including measuring the current-voltage offset value andcomparing the measured offset value to a capacitance mode detectionthreshold value.

In step 1406, if the system determines that the source resonator isoperating in a capacitive mode, then control passes to step 1408, wherethe system reduces the output power of the source resonator to ensurethat failure of resonator components such as inverter switches does notoccur. Control then returns to step 1404, where the system determineswhether the source resonator is still operating in a capacitive mode atlower output power.

If the system determines that the source resonator is not operating in acapacitive mode, then control passes to optional step 1410, where thesystem determines whether the source resonator is operating in areactive mode. Some or all of the steps disclosed in connection withFIG. 9 can be used to make this determination, including measuring thecurrent-voltage offset value, measuring the output current at thevoltage switching point, and measuring the bus voltage, and using themeasured values of these operating parameters to determine whether thesource resonator is operating in a reactive mode.

In step 1412, if the system determines that the source resonator isoperating in a reactive mode, then control passes to step 141, where thesystem reduces the output power of the source resonator so that theresonator represents a smaller inductive load. Control then returns tostep 1404, where the system determines whether the source resonatoroperates in a capacitive mode (and a reactive mode) at lower outputpower.

If the system determines that the source resonator is not operating in areactive mode (nor in a capacitive mode), then in step 1416, the systemdetermines the efficiency of power transfer between the source resonatorand the receiver resonator. The efficiency can be determined in variousways. In some embodiments, for example, the receiver resonator transmitsa signal to an electronic processor or the system that includesinformation about the amount of power received by the receiverresonator. The electronic processor can use this information, along withinformation about the output power of the source resonator, to determinean efficiency of power transfer.

In step 1418, the efficiency of power transfer is compared to a transferefficiency threshold value. If the efficiency of power transfer isgreater than the threshold value, then power transfer is occurring at asuitable efficiency, and the procedure terminates at step 1422. If theefficiency of power transfer is smaller than the threshold value, thenthe impedance of the source resonator is adjusted in step 1420 (e.g., tobetter match a target impedance value that leads to greater powertransfer efficiency) to improve the power transfer efficiency. Controlthen returns to step 1404, and the system determines whether the sourceresonator is operating in a capacitive or reactive mode at the adjustedimpedance value.

In some embodiments, the target impedance value discussed herein isknown beforehand from design specifications and/or calibration of thewireless power transfer system for a variety of operating conditions andgeometries. In certain embodiments, the target impedance value is notknown, and is instead achieved by iteratively adjusting the system basedon feedback criteria such as power transfer efficiency. In eithercircumstance, the impedance of the system is adjusted to match thetarget impedance for efficient transfer of power from the source (e.g.,a power amplifier) to a receiver (e.g., a receiver resonator).

Mode Detectors

In the foregoing discussion, a mode detector is used to measureoperating parameters such as the current-voltage offset value to allowthe system to determine whether the source resonator is operating in acapacitive mode or a reactive mode. In general, a variety of differentmode detectors can be used for this purpose, and in principle, any modedetector that provides an accurate, rapid measurement of thecurrent-voltage offset can be used.

FIG. 11 shows a schematic diagram of one example of a mode detector1500. Mode detector 1500 includes a signal processing unit 1502 and anoffset measurement unit 1504. Signal processing unit 1502 includes afirst input terminal that receives a current waveform from a detector orcircuit such as a zero crossing detector (ZCD), and a second inputterminal that receives a voltage waveform from a detector or circuitsuch as a pulse-width modulation (PWM) detector. The current and voltagewaveforms are provided to an AND logic gate 1506, which produces anoutput signal that is provided to one input terminal of an XOR logicgate 1508. The voltage waveform is provided to the other input terminalof XOR gate 1508. The output waveform 1510 from XOR gate 1508 is apositive-only square waveform with a temporal width 1512 thatcorresponds to a temporal offset between the current and voltagewaveforms that are provided to the input terminals of AND gate 1506.

Waveform 1510 is provided to one input terminal of offset measurementunit 1504. A clock signal is provided to another input terminal ofoffset measurement unit 1504. The clock signal consists of a series ofpulses spaced at a regular temporal interval, and can be provided by asignal generator, by a microcontroller such as microcontroller 1514, byother logic units or circuits within the system, or by a systemcontroller.

Offset measurement unit 1504 implements an internal counter CNT thatincrements as each additional clock signal pulse is received at thesecond input terminal of offset measurement unit 1504. Waveform 1510functions as the on-off gate signal during which counter CNT isincremented. To measure the temporal offset between the current andvoltage waveforms, which corresponds to the temporal width 1512 ofwaveform 1510, the value of CNT is initially reset to zero by offsetmeasurement unit 1504. The value of CNT remains at zero until theleading edge of waveform 1510 is detected at the first input terminal ofoffset measurement unit 1504, at which time measurement of the width ofwaveform 1510 is initiated. During measurement of the width of waveform1510, each time a clock signal pulse is detected at the second inputterminal of offset measurement unit 1504, the value of CNT isincremented. Accumulation in CNT continues until the trailing edge ofwaveform 1510 is detected at the first input terminal, at which timemeasurement of the width of waveform 1510 terminates. The value of CNT,which corresponds to the temporal offset between the current and voltagewaveforms in units of clock cycles, is provided at the output terminalof offset measurement unit 1504. The temporal offset in units of timecan be obtained by performing a subsequent multiplication operation(e.g., within offset measurement unit 1504 or in another logic unit suchas an electronic processor) in which CNT is multiplied by the constanttemporal interval between clock cycles.

In general, mode detector 1500 is positioned within the source resonator(e.g., power amplifier). A variety of different arrangements of the modedetector within the source resonator can be used. FIG. 12 is a schematicdiagram of an example of a power amplifier 1600 that includes modedetector 1500. In FIG. 12, an integrated controller 1620 controls avariety of different functions of amplifier 1600. In particular,controller 1620 adjusts the impedance of source coil 1640 usingimpedance matching network 1630. Zero crossing detector 1650 measuresthe current waveform corresponding to the output current, and a pulsewave modulation detector (indicated schematically as “PWM”) measures thevoltage waveform. The current and voltage waveforms are provided toinput terminals of mode detector 1500. The AND and XOR logic gates, 1506and 1508 respectively, are shown in FIG. 12.

In FIG. 11, output waveform 1510 from XOR gate 1508 is provided tooffset measurement unit 1504, which is part of mode detector 1500. Insome embodiments, however, mode detector 1500 does not include an offsetmeasurement unit. FIG. 12 shows such an embodiment. In FIG. 12, outputwaveform 1510 is provided directly to integrated controller 1620.Integrated controller 1620 also receives a clock signal (not shown inFIG. 12) and uses output waveform 1510 as a gate signal to measure thetemporal offset between the current and voltage waveforms measured byzero crossing detector 1650 and the PWM detector, respectively, in amanner similar to the procedure described above for offset measurementunit 1504.

As shown in FIG. 12, amplifier 1600 can also include a current sensor1610. Sensor 1610 receives, on two input terminals, the current waveformmeasured by zero crossing detector 1650, and the voltage waveformmeasured by the PWM detector. Current sensor 1610 produces an outputsignal 1660 that corresponds to the output current at the voltageswitching point. This signal is provided to integrated controller 1620.

FIG. 12 shows one example of the measurement of current and voltagesignals to provide input waveforms to mode detector 1500. Current and/orvoltage signals can also be measured at other points in a sourceresonator to provide input waveforms to a mode detector. For example, inboth the single-ended and differential amplifier topologies (shown inFIGS. 3 and 4, respectively), the input current to the impedancematching networks 342 driving the resonator coils 344 can be obtained bymeasuring the voltage across a capacitor 324, or via a current sensor ofsome type. For the exemplary single-ended amplifier topology in FIG. 3,the current may be sensed on the ground return path from the impedancematching network 342. For the exemplary differential power amplifierdepicted in FIG. 4, the input current to the impedance matching networks342 driving the resonator coils 344 may be measured using a differentialamplifier across the terminals of capacitor 324 or via a current sensorof some type. In the differential topology of FIG. 4, capacitor 324 canbe duplicated at the negative output terminal of the source poweramplifier.

In both topologies, after single-ended signals representing the inputvoltage and current to the source resonator and impedance matchingnetwork are obtained, the signals can be filtered to obtain desiredportions of the signal waveforms. For example, the signals can befiltered to obtain the fundamental component of the signals. The typesof filtering that can be performed include, for example, low pass,bandpass, and notch. Filter topologies can include elliptical,Chebyshev, and Butterworth, for example.

Tunable Circuit Elements

A variety of different tunable circuit elements can be used in impedancematching networks to adjust the impedance of a source and/or receiverresonator. FIG. 13 is a schematic diagram showing a portion of awireless power transfer system 3000. System 3000 includes a powertransmitting apparatus and a power receiving apparatus. In FIG. 13,portions of the power transmitting apparatus and the power receivingapparatus are schematically drawn as a source-side circuit 3002 and areceiver side circuit 3032, respectively.

The source-side circuit 3002 can include an inductor 3004 and capacitors3005-3008. The inductance of inductor 3004 can correspond to theinductance of a source resonator of the power transmitting apparatus insystem 3000. In some embodiments, any of the capacitors 3005-3008 can bevariable capacitors which are used for tuning an impedance of thesource-side circuit 3002.

In certain embodiments, the source-side circuit 3002 can also includetunable inductors 3010 and 3014. Tunable inductor 3010 is connected tothe source-side circuit 3002 by terminals 3011 and 3012 of the tunableinductor. Tunable inductor 3014 is connected to the source-side circuit3002 by terminals 3015 and 3016 of the tunable inductor. In the exampleshown in FIG. 13, the tunable inductors 3010 and 3014 are connected tothe source-side circuit 3002 in a series connection. In otherembodiments, tunable inductors 3010 and 3014 can be connected to thesource-side circuit 3002 in a parallel connection.

The receiver-side circuit 3032 can include inductor 3034 and capacitors3035-3036. The inductance of inductor 3034 can correspond to inductanceof a receiver resonator of the power receiving apparatus in system 3000.In some embodiments, any of the capacitors 3035-3036 can be variablecapacitors used for tuning an impedance of the receiver-side circuit3032. In certain embodiments, the receiver-side circuit 3032 can includea tunable inductor 3040, which is used to tune the impedance of thereceiver side circuit 3032.

Terminals 3020 and 3021 of the source-side circuit 3002 can be connectedto a power circuit (not shown) so that the source-side circuit 3002 canreceive AC voltages and currents from the power circuit. In someembodiments, the power circuit can include a power amplifier. Thereceived AC voltages and currents can generate alternatingelectromagnetic fields via inductor 3004, where the alternating fieldsare used to transfer power to the receiver-side circuit 3032 through theinductor 3034.

Generally, the source-side circuit 3002 and the receiver-side circuit3032 can include resistors and other circuit elements, which are notshown in FIG. 13 for clarity. Furthermore, although one arrangement ofcapacitors and inductors in source-side circuit 3002 and receiver-sidecircuit 3032 is shown in FIG. 13, more generally, both source-sidecircuit 3002 and receiver-side circuit 3032 can include otherarrangements of capacitors and inductors.

While both tunable capacitors and tunable inductors can be used toadjust the impedance of source and receiver resonators in the systemsand methods disclosed herein, tunable inductors can provide certainadvantages in high power applications. For example, when large amountsof power are transferred wirelessly, tuning impedance using capacitorscan involve switching among banks of capacitors to provide suitabletuning ranges. However, switching among banks of capacitors frequentlyleads to power losses that are not insignificant. For low powerapplications, such power losses can be tolerable. For high powerapplications, such power losses can be more important, as they have agreater effect on the amount of time required for delivery of power. Asan example, switching among banks of capacitors may cause power lossesthat are unacceptable when delivering power for charging vehiclebatteries.

Switching among capacitor banks at high power levels can also lead toinefficient coupling. In particular, it can be difficult to maintainhigh Q operation of source and receiver resonators and, at the sametime, operate under high voltage and current levels, when switchingamong capacitor banks to provide impedance tuning. Switches used forselecting different capacitor banks can also be prone to failure at highpower transfer rates.

Tunable inductors provide an alternative to tunable capacitors forpurposes of impedance matching in wireless power transfer systems.Moreover, certain types of tunable inductors, due to their structures,are particularly well suited for operation at high power transfer rates.These inductors are therefore well suited for high power applicationssuch as vehicle charging, where tunable capacitors and other types oftunable inductors have certain shortcomings. In the following section, avariety of different tunable inductors are disclosed, each of which issuitable for inclusion in impedance matching circuits and networks, andeach of which is particularly well suited to high power applicationsinvolving wireless power transfer.

FIGS. 14A and 14B are schematic diagrams showing an example of a tunableinductor 3100, which can be used as tunable inductors 3010 and 3014described in relation to FIG. 13. For example, each of the tunableinductors 3010 and 3014 can correspond to tunable inductor 3100.

In the example described in FIGS. 14A and 14B, the inductance of thetunable inductor 3100 can be electrically tuned over a wide range. FIG.14A schematically depicts the structure of magnetic materials of thetunable inductor 3100. Other elements (e.g., conducting wire) of thetunable inductor 3100 are not shown. In this example, the tunableinductor 3100 includes two magnetic materials 3102 and 3104. Each of themagnetic materials 3102 and 3104 has an “E-shaped” structure. Themagnetic material 3102 includes portions 3110, 3112 and 3114, whichprotrude as “legs” from base 3103 of the magnetic material 3102 so thatthe magnetic material 3102 has an E-shape. Similarly, the magneticmaterial 3104 includes portions 3120, 3122 and 3124, which protrude aslegs from base 3105 of the magnetic material 3104.

The portions 3110 and 3120 face each other to form gap 3130. Similarly,portions 3112 and 3122 face each other to form gap 3132, and portions3114 and 3124 face each other to form gap 3134. In some embodiments, anyof the gaps 3112, 3132 and 3134 can include air to form an “air gap.” Incertain embodiments, any of the gaps 3112, 3132 and 3134 can include oneor more dielectric materials such as paper. The gaps act as an impedanceto magnetic field flux generated within the magnetic materials 3102 and3104. Such impedance can affect the inductance value of the tunableinductor 3100. Thus, the type of material provided within gaps 3112,3132 and 3134 can be selected based on the desired impedance value. Forexample, the material can be selected to provide an inductance valuecorresponding to a value in the middle of the range of the tunableinductance range for inductor 3100.

In some embodiments, tunable inductor 3100 can include a support to holdmagnetic materials 3102 and 3104 relative to each other with gaps 3130,3132 and 3134. For example, a support structure (not shown) can be usedto hold the legs protruding from base 3103 and 3105 while maintainingthe gaps 3130, 3132 and 3134.

FIG. 14B is a schematic diagram of another view of tunable inductor 3100shown in FIG. 14A. The tunable inductor 3100 includes a coil 3150, whichis wound around portions 3112, 3122 and gap 3132. When tunable inductor3100 is introduced into the source-side circuit 3002 of FIG. 13,terminals 3151 and 3152 of coil 3150 can be connected to terminals 3011and 3012, respectively, or to terminals 3015 and 3016, respectively.Accordingly, the inductance of coil 3150 in the tunable inductor 3100can correspond to the inductance value of the tunable inductor 3010 orthat of the tunable inductor 3014.

Referring to FIG. 14B, a control circuit 3140 is connected to coil 3142,which may be referred as an “excitation coil” in this disclosure. Thecoil 3142 is wound around portions 3110, 3120 and gap 3130. In thisexample, the coil 3142 is further wound around portions 3114, 3124 andgap 3134, and then connected to the control circuit 3140. The controlcircuit 3140 can be configured to provide a DC bias on the magneticmaterials 3102 and 3014 by sending a DC current through coil 3142. Forexample, the control circuit 3140 can send DC current 3102 in thedirection shown in FIG. 14B, at a given time. The DC current 3102flowing through coil 3142 generates magnetic fields in directions 3175,which change an effective permeability of the magnetic materials 3102and 3104 at the operating frequency of power transfer. Because theinductance of the coil 3150, which is wound around portions 3112, 3122and gap 3132, depends on the effective permeability of the magneticmaterials 3102 and 3104, a change in the effective permeability leads toa change of the inductance of the coil 3150 in the tunable inductor3100. The magnitude of changes of the effective permeability and theinductance depends on the magnitude of the DC current 3102. Thus,current 3102 can be used to magnetize the magnetic materials 3102 and3104 and thereby to adjust inductance of tunable inductor 3100. Themagnitude of current 3102 can be determined from change in inductancedesired to achieve a particular rate of power transfer, given factorssuch as the relative positions of the source and receiver resonators, asdiscussed previously.

Moreover, because the coil 3142 is also wound around portions 3114, 3124and gap 3134, the change of effective permeability and the inductancecan be twice compared to the case where the coil 3142 is wound onlyaround portions 3112, 3122 and gap 3132. In some embodiments, thevoltage applied to coil 3142 can be clamped or regulated to control thechange of inductance. In certain embodiments, coil 3142 can includemultiple taps to provide higher resolution of tuning and lesser currentlevels.

In general, the size of gaps 3130, 3132, and 3134 are selected based onthe current levels within the windings and the saturation of materials3102 and 3104. In some embodiments, for example, the gaps are between 1mm and 10 mm.

The tunable inductors 3010 and 3104 can be connected in series orparallel to the source-side circuit. The control circuit 3140 can beconfigured to electrically tune the inductances of the tunable inductors3010 and 3014 at the same time. Phases of voltages applied to thetunable inductors 3010 and 3014 for tuning can be arranged such thatvoltage across coil 3142 cancels to minimize the voltage ratings of thecontrol circuit 3140.

In some embodiments, the control circuit 3140 can be connected to thecoil 3142 by way of a shorting switch. When tuning is not needed, theswitch can be opened so that control circuit 3140 does not pass currentthrough coil 3142. When tuning is needed, the switch can be closed andthe control circuit 3130 can provide current through coil 3142.

FIG. 14C is a schematic plot 3180 showing the inductance of the coil3150 as a function of H (amperes-turn/cm) of the example shown in FIG.14B. In this example, H can be expressed as H=(2NI)/L, where N is thenumber of turns wound around portions 3110, 3120 and gap 3130, and L isthe length of the magnetic flux path. I is the magnitude of current 3102and L is the length 3170 of the winding of coil 3142. The factor of 2arises due to the fact that the number of turns N of coil 3142 is thesame for portions 3110, 3120, gap 3130 and portions 3114, 3124, gap3134. In some embodiments, the number of turns N can be 9 or more (e.g.,15 or more, 25 or more, 35 or more, 45 or more). The ratio of number ofturns N to the winding turns of coil 3150 can be 1:1. In certainembodiments, a larger number of turns N can lead to a larger couplingbetween coil 3142 and 3150 which can provide a larger inductance changecompared to embodiments with fewer turns.

Axis 3182 of plot 3180 is H (amperes-turn/cm). Axis 3184 of plot 3180 isthe inductance of coil 3150. Below threshold 3190 of H, the inductancecan be substantially flat as shown in line 3186 compared to line 3188above the threshold 3190. A linear dependence of line 3188 indicatesthat the inductance of coil 3150 and power can be adjusted linearly.More generally, for certain tunable inductors, lines 3186 and 3188 maynot be perfectly straight, but can instead be curved.

As illustrated in the FIG. 14C, the inductance of coil 3150 can be tunedby adjusting a magnitude of current 3102 (I) using the control circuit3140. In particular, the inductance can be effectively tuned abovethreshold 3190. In this approach, the inductance of the tunable inductor3100 can be electrically tuned by adjusting the amount of currentprovided by the control circuit 3140.

Referring back to FIG. 13, system 3000 includes two tunable inductors3010 and 3014. In some embodiments, the inductances of the tunableinductors 3010 and 3014 can be tuned so that their respective inductancevalues are substantially the same. In certain embodiments, theinductances of the tunable inductors 3010 and 3014 can be tuned so thattheir respective inductance values are within 1% (e.g., within 3%,within 5%, within 10%) of one another with respect to the larger value.By having the inductance values of the tunable inductors 3010 and 3014to be substantially the same, the impedance tuning of the source-sidecircuit 3002 can be balanced. Such balance can reduce electromagneticinterference (EMI) generated by the source-side circuit 3002 becauseelectromagnetic induction or electromagnetic radiation generated bytunable inductors 3010 and 3014 may cancel out each other. The EMI mayaffect wireless communication between the source-side circuit 3002 andreceiver-side circuit 3032.

In some embodiments, the inductances of tunable inductors 3010 and 3014can be tuned using separate control circuits 3140. Alternatively, insome other embodiments, a single control circuit 3140 can connect totunable inductors 3010 and 3014. For example, the single control circuit3140 can provide currents flowing through a coil 3142 in the tunableinductor 3010 along to a coil 3142 in the tunable inductor 3014. In thisapproach, one end of coil 3142 in the tunable inductor 3010 is connectedto one end of coil 3142 of the tunable inductor 3014. This approach canbe advantageous by using one control circuit 3140 to control theinductance of both tunable inductors 3010 and 3014. Because the currentflowing through tunable inductors 3010 and 3014 may be the same, thechanges of the inductances of the tunable inductors 3010 and 3014 can bethe same, and balanced tuning can be achieved.

Balanced tuning of both inductors using a single control winding inseries and a single control circuit 3140 can be particularlyadvantageous for certain applications. In particular, by using the samecurrent excitation in both inductors while the inductors are tuned,balanced tuning of the inductors is guaranteed, without the possibilityof a mismatch in inductance during tuning.

In certain embodiments, the control circuit 3140 can also send DCcurrent 3102 in an opposite direction than shown in FIG. 14B. Suchbi-polar direction of DC current 3102 may provide a wider range oftuning of inductance compared to the case when DC current is sent inonly one direction.

Generally, tunable inductors 3010 and 3014 can be used in thereceiver-side circuit 3032 to tune impedance of the receiver-sidecircuit 3032 in a balanced manner.

FIGS. 15A and 15B are schematic diagrams showing an example of magneticmaterial having an E-shaped structure, which can be used in the tunableinductor 3100 described in relation to FIGS. 14A-C. In some embodiments,dimensions 3201-3208 can be 56.2 mm, 27.6 mm, 24.61 mm, 18.5 mm, 37.5mm, 17.2 mm, 9.35 mm and 10.15 mm, respectively. Dimension 3203 is thethickness of a magnetic material as indicated in FIG. 15A. In someembodiments, dimensions 3201-3208 can be 56.1 mm, 23.6 mm, 18.8 mm, 14.6mm, 38.1 mm, 18.8 mm, 9.5 mm and 9.03 mm, respectively. The magneticmaterial can include ferrites such as Fe and Ni, and other soft ferritematerials that are designed for power applications. The magneticmaterial can be used for wireless power transfer of 1 kW or more (e.g.,2 kW or more, 3.3 kW or more, 4.5 kW or more, 5.5 kW or more, 6.4 kW ormore).

FIG. 16A is another schematic diagram showing power transfer system 3000described in relation to FIG. 13. In FIG. 16A, some of the capacitors(e.g., variable capacitors) and inductors of the system 3000 areschematically depicted as matching circuit 3304 and 3302. Source-sidecircuit 3002 is connected to a power circuit 3302 which can include aswitch-mode power supply for generating alternating electromagneticfields through inductor 3004. The power circuit 3302 can be connected totunable inductors 3010 and 3014 by terminals 3311 and 3312,respectively. In this disclosure, the voltage 3314 across terminals 3315and 3316 of load 3050 can be referred to as a load voltage (Vload).Further, the bus voltage (Vbus) is typically the input voltage to theamplifier or the voltage that is generated from the power factorcorrector, while the root-mean-square voltage 3310 across terminals 3311and 3312 is the output voltage of the amplifier.

FIG. 16B is a schematic diagram showing a portion of the power transfersystem 3000 depicted in FIG. 16A. For the source-side circuit 3002,inductor 3004 (e.g., a coil) having terminals 3320 and 3321 is woundaround a magnetic material 3330. For the receiver-side circuit 3032,inductor 3034 (e.g., a coil) having terminals 3315 and 3316 is woundaround a magnetic material 3332. In this example, magnetic materials3330 and 3332 are planar magnetic materials, and inductors 3004 and 3035can generate magnetic dipole moments in a plane of the planar magneticmaterials, respectively. In this disclosure, distance 3336 between themagnetic materials 3330 and 3332 is referred as a “z distance.”

FIGS. 17A-D are images showing an example of the operation of the system3000, in which the inductance of tunable inductors 3010 and 3014 istuned to adjust impedance matching conditions; distance 3336 is 12 cm.In one measurement, the power circuit 3302 delivered 2.5 kW to the load3050 while adjusting voltage 3314. The adjustment of voltage 3314changed the impedance of the receiver-side circuit 3032. For example,such changes can simulate voltage changes of a battery load when beingcharged over time. In this measurement, the voltage 3314 was adjustedbetween 250 V and 360 V while operating the load 3050 in voltage moderather than in current mode.

Tunable inductors 3010 and 3014 were electrically tuned to optimize theimpedance matching between source-side circuit 3002 and receiver-sidecircuit 3032 while the impedance of the receiver-side circuit 3032changed due to the adjustment of voltage 3314. The tuning of inductors3010 and 3014 allowed power transfer to be about 2.5 kW±10% for therange of 240 V-380 V of voltage 3314. In contrast, without tuninginductances, the load voltage range over which approximately constantpower delivery is possible is approximately 60 V. Accordingly, the rangeof voltages 3314 over which a substantial change in power transfer ratedid not occur (e.g., the power transfer rate was maintained at 2.5kW±10%) was larger by a factor of about 2.5 than the range of voltagesover which the power transfer rate would have been maintained withouttuning the tunable inductors 3010 and 3014.

In another measurement, power transfer of 3 kW with less than 10%variation was achieved when adjusting voltage 3314 between 240 V and 380V. The voltage 3310 was about 290 V. Voltage 3310 can be increased byincreasing the input impedance of the tunable inductors 3010 and 3014 sothat they will operate within the operating range of the power factorcorrector (PFC).

FIG. 17A is an image showing the measured parameters of the system 3000during the foregoing measurement. Display 3402, which is labeled as“Urms1,” indicates the root-mean-square voltage 3310 across terminals3311 and 3312 of power circuit 3302. Display 3404, which is labeled as“Irms1,” indicates root-mean-square current passing terminal 3311. Sucha current can be measured by a current sensor. Display 3406, which islabeled as “P1,” indicates the power provided by the power circuit 3302.Display 3408, which is labeled as “S1,” indicates the a voltage-currentproduct (e.g., 4196 VA). For an AC source, the voltage-current productis a measure of how much power a source has to provide for a given load.Display 3410, which is labeled as “P2,” indicates the power received byload 3050. Display 3412, which is labeled as “λ1,” indicates the powerfactor. Display 3414, which is labeled as “η1,” indicates the percentageratio of P2 to P1. Display 3416, which is labeled as “Udc2,” indicatesvoltage 3314 across load 3050.

In the measurement shown in FIG. 17A, control circuit 3040 did not applycurrent to tunable inductors 3010 and 3014. FIG. 17B is an image 3320showing observed voltages and currents using an oscilloscope during thismeasurement. Trace 3422 indicates the trace of voltage 3422, and trace3424 indicates the trace of current Irms1 displayed by display 3404. Thephase difference 3450 between trace 3422 and trace 3424 indicates thatsystem 3000 is operating close to capacitive operation. Trace 3426indicates the trace of current provided by control circuit 3040 to thetunable inductors 3010 and 3014. Trace 3428 indicates the trace ofvoltage provided by control circuit 3040 to the tunable inductors 3010and 3014.

FIG. 17C is an image 3330 showing measured parameters of the system 3000in another operating condition. The power provided by the power circuit3302 was increased to about 3.6 kW compared to that of the operationshown in FIGS. 17A and 17B. Accordingly, display 3406 indicates thepower provided by the power circuit 3302 to be about 3.6 kW. Display3410 indicates the power received by load 3050 to be about 3.2 kW.Display 3416 indicates voltage 3314 across load 3050 to be about 380 V.During this operation, control circuit 3040 applied currents to thetunable inductors 3010 and 3014 to electrically tuning the inductancesof the tunable inductors 3010 and 3014. The tunable inductors 3010 and3014 were tuned to optimize the impedance matching between source-sidecircuit 3002 and receiver-side circuit 3032 so that of display 3414 wasmaintained to be about 90% while the P1 was changed from 3 kW to 3.6 kWand Udc2 was changed from 240 V to 380 V. These measurement resultsdemonstrated that impedance changes due to changes in Udc2 can bematched by tuning the tunable inductors 3010 and 3014. The voltage 3314presented by Udc2 was extended by 140V without adjusting voltage 3310 bymore than 5%. In certain embodiments, the voltage 3310 can be adjustedto provide an increase of voltage 3314.

In the measurement shown in FIG. 17C, control circuit 3040 appliedcurrents to tunable inductors 3010 and 3014 for tuning the inductance ofthe inductors. FIG. 17D is an image 3440 showing observed voltages andcurrents using an oscilloscope during this measurement. Phase difference3452 between trace 3422 and trace 3424 indicates that while theamplifier load was inductive, it was tuned to be less inductive than itotherwise would have been without tuning. Trace 3426 indicates thatcurrents were provided by control circuit 3040 to the tunable inductors3010 and 3014. The currents were about 3 A. The currents can be reducedby increasing the number of turns N of the excitation coil. Trace 3428was close to zero. In the above measurements, when current 3102 wasvaried from 0 to 4 A, the inductance of tunable inductors 3010 and 3104varied from about 33 pH to about 22 μH.

The measurements described in relation to FIGS. 17A-D demonstrate thathigh power transfer can be achieved for a wide range of load voltages3314 by tuning the inductance of tunable inductors 3310 and 3314. Themeasurement results were achieved without varying the operationalfrequency of power transfer. In some embodiments, control circuit 3040can be a simple circuit to provide adjustable currents or voltages toexcitation windings of the tunable inductors 3310 and 3314. A singlecontrol circuit 3040 can tune both tunable inductors 3310 and 3314. Incertain embodiments, the control circuit 3040 can tune the tunableinductors 3310 and 3314 when impedance matching conditions change due toa change in relative position (e.g., x-offset, y-offset, z-offset)and/or orientation (rotation, tilt) between inductors 3004 and 3034.

In certain embodiments, variable frequency operation, where theoperational frequency of the power transfer is adjusted, can be utilizedalong with tuning of tunable inductors in system 3000. The aboveapproaches may be implemented along with adjusting Vbus so that a widerange (e.g., 180 V, 200 V) of load voltages is acceptable. In addition,the above approaches can be implemented without using special magneticmaterials, which are used for magnetic amplifiers. In some embodiments,magnetic materials such as ferrites, including iron oxide, can be used.

FIG. 18 is a schematic diagram of another example of a tunable inductor3500 that can be used for tunable inductors 3010 and 3014 describedabove. In this example, the tunable inductor 3500 includes a magneticmaterial 3502 which is shaped as two E-shaped portions facing eachother. The magnetic material 3502 include leg portions 3504 and 3506facing each other and forming a gap 3508. Similar to tunable inductor3100 described in FIG. 14B, coil 3150 (not shown) is wound aroundportions 3504, 3506 and gap 3508. Coil 3142 (not shown), which isconnected to a control circuit 3140 (not shown), is wound aroundportions 3510 and 3520, in a manner similar to tunable inductor 3100.

In some embodiments, a tunable inductor can include a toroid-shapedmagnetic material with a gap (e.g., an air gap). A coil 3150 can bewound around the gap in a similar while a coil 3142 can be wound aroundthe toroid-shaped magnetic material to adjust its effective permeabilityin a similar manner described in relation to FIG. 14B.

The tunable inductors 3010 and 3014 can have smaller inductance valuesthan that of matching circuit 3304. In some embodiments, additionalsmall inductors can be connected in series to the tunable inductors 3010and 3014 for impedance matching.

In certain embodiments, electrically tuning the inductance values can beadvantageous over mechanically tuning the inductance values becausemechanically tuning can be more vulnerable to vibration andcontamination, and thus be less robust in certain applications. Inaddition, for mechanical tuning, a magnetic flux variation in a gapregion of a magnetic material may reduce performance. For example, whenthe gap size is mechanically varied to have a larger size, turnings ofthe wire may provide a significant magnetic flux so as to generateheating of the wire and magnetic material. The heating can lead toshorting or damage of the tunable inductor.

Nonetheless, in some embodiments, mechanically-tuned inductors can beused for impedance matching in wireless power transfer systems. FIGS.19A and 19B are schematic diagrams of another example of a tunableinductor 3600, which can be used for tunable inductors 3010 and 3014.Referring to FIG. 19A, the tunable inductor 3600 can include a coil 3608wound around a magnetic material 3602. The magnetic material includestwo portions 3603 and 3604 which are displaced from one another to forma gap 3605. In this example, the gap includes air to form an air gap. InFIG. 19A, gap 3605 has a separation distance 3638. One end of coil 3608can be connected to, for example, terminal 3011 and the other end toterminal 3012 shown in FIG. 13. At a given time, currents in portion3609 of the coil 3608 can flow in a direction outwards of the drawingplane (as indicated by the “dots”) and currents in portion 3610 can flowin a direction inwards to the drawing plane (as indicated by the“crosses”). The currents can be passed on to a source resonator insource-side circuit 3002 for power transfer.

The tunable inductor 3600 can include an actuator 3619 used to displaceportion 3604 of the magnetic material 3602, and thereby changing thedistance 3638 of the gap 3605. Such a change leads to a change of theeffective permeability of the magnetic material 3602 seen by the coil3608. The change of effective permeability leads to a change of theinductance of the coil 3608. Accordingly, the inductance of the tunableinductor 3600 can be tuned by mechanically varying the distance 3638.

The actuator 3619 can include a deformable element 3621. For example,the deformable element 3621 can be a shape memory alloy (SMA), which canchange its shape when a current is passed through it. In someembodiments, currents passing through the shape memory alloy heat theshape memory alloy, thereby deforming its shape. In certain embodiments,the deformable element 3621 can be an SMA wire or an SMA foil. The SMAcan include Nitinol (nickel titanium) material, copper-aluminum-nickel,or alloys of zinc, copper, gold and iron, for example.

In some embodiments, the deformable element 3621 can have a springstructure as illustrated in FIG. 19A. One end of the deformable element3621 is fixed onto a support structure 3620 and the other end of thedeformable element 3621 is fixed onto a support structure 3622. Thesupport structure 3622 can guided through base 3624 along direction3628. In the example illustrated in FIG. 19A, the actuator 3619 includesa bias spring 3630, where one end is fixed to a holding element 3625(e.g., a plastic cap). The holding element 3625 is configured to moveportion 3604 by displacing fixture element 3631 fixed to portion 3604and holding element 3625 on each end, respectively.

The deformable element 3621 can apply force to support structure 3622 indirection 3634. On the other hand, the bias spring 3630 can apply forceto holding element 3625 and the support structure 3622 in direction3632. The forces applied on each end of the support structure 3622 canbalance out each other so that fixture element 3631 can hold the portion3064 in a fixed position. In some embodiments, the deformable element3621 can be deformed, for example, by passing currents through thedeformable element. For example, the deformable element 3621 can shrinkwhen heated by the currents. In another example, the deformable element3621 can expand when heated by the currents. In these approaches, thedeformable element 3621 can apply varying forces onto the supportstructure 3622.

FIG. 19B is a schematic diagram of the tunable inductor 3600 shown inFIG. 19A after applying currents through the deformable element 3621. Inthis example, the deformable element 3621 has contracted so thatdistance 3638 decreased. As a result, the inductance of the tunableinductor 3600 has varied. Length 3639 of the deformable element 3621 canbe controlled by adjusting the magnitude and duration of current appliedthrough the deformable element 3621. In this approach, the inductance ofthe tunable inductor 3600 can be tuned by a user by adjusting the length3639.

In certain embodiments, the tunable inductor 3600 can include multiplebias springs. Alternatively, in some embodiments, the tunable inductor3600 may not include a bias spring and the deformable element 3621 canadjust the distance 3638 without using a bias spring.

Generally, one or more deformable elements 3621 can be included in asingle actuator.

FIG. 20 is an image 3700 of an example of a tunable inductor 3701, whichincludes magnetic material 3602. The magnetic material 3602 includes twoportions 3603 and 3604 where a gap 3605 is formed between the portions3602 and 3604. In this example, each of the two portions 3602 and 3604form an E-shaped structure. The two portions 3602 and 3604 are held bysupport structure 3702, which includes a spring (not shown) so thatlength 3708 is variable. A coil is wound around the center legs of theE-shaped structures of portions 3602 and 3604 in a manner similar tocoil 3150 described in FIG. 14B. The coil and the center legs arecontained within the support structure 3702. One end of the coil isconnected to electrode 3710 and the other end to the electrode 3711 sothat currents can pass through the coil by way of the electrodes 3710and 3711. In some embodiments, the tunable inductor 3701 can be used ineither a source-side circuit or a receiver-side circuit so that the coilcan acts as a tunable inductor. For example, a power circuit can providecurrents through electrodes 3710 and 3711 and then to a source resonatorincluded in the source-side circuit.

In some embodiments, the tunable inductor 3701 can include electrodes3704 and 3706. A control circuit can be used to apply currents fromelectrodes 3704 to electrodes 3706 by way of deformable element 3621. Inthis example, the currents can heat the deformable element 3621, and theheating can change the length of the deformable element 3621 so that theseparation between the two portions 3602 and 3604 can be adjusted. Asmentioned earlier, the adjustable separation can lead to tunableinductance values of the tunable inductor 3701. In this approach, theinductance of tunable inductor 3701 can be mechanically adjusted byapplying electrical currents provided by the control circuit.

In some embodiments, the deformable element 3621 can be deformed byapplying a voltage to an adjacent heating or cooling element, whichcontrols the temperature of the deformable element 3621.

In the foregoing examples of tunable inductors, the magnetic materialcan include hard ferrite or soft ferrites. The magnetic material caninclude Fe or other ferrous material. In certain embodiments, themagnetic material can include magnetic powdered cores made from Ni, Feand Mo. Coils can include solid or hollow wires made from materials suchas copper, aluminum. In some embodiments, a coil can include a Litzwire. For example, magnetic powdered cores and Litz wire can be utilizedat an operating frequency of 85 kHz for wireless power transfer, due totheir low loss.

In some embodiments, one or more tunable elements with capacitancetuning capabilities can be used for impedance tuning in a wireless powertransfer system. Such tunable elements can tune an impedance of acircuit of the system by tuning capacitance values instead of inductancevalues. Such tunable elements can be used instead of tunable inductors3010 and 3014 in power transfer systems 3000 and 3300.

FIG. 21 is a schematic diagram of a portion 3800 of a power transfersystem including an example of a tunable element 3802 with capacitancetuning capabilities. The portion 3800 can correspond to a portion of asource-side circuit or a receiver-side circuit. In this example, theportion 3800 corresponds to a portion of the source-side circuit. Theportion 3800 can include capacitors 3604, 3606 and 3608 which can havefixed value of capacitances or can be variable capacitors to provide adesired impedance value of the source-side circuit.

The tunable element 3802 can include a p-n junction between twodifferent types of a semiconductor material. The capacitance of thetunable element 3802 can be tuned by adjusting a nonlinear depletioncapacitance of the reverse biased p-n junction. For example, to achievethis, the tunable element 3802 can include two diodes 3810 and 3811connected back-to-back as shown in FIG. 21, where each diode includes ap-n junction. Anodes of the diodes 3810 and 3811 are connected to eachother. In this arrangement, capacitance of the combined diodes 3810 and3811 depends on the applied voltage across the tunable element 3802.When an AC voltage is applied to the tunable element 3802, the diodes3810 and 3811 act as rectifiers charging their depletion capacitances toa reverse bias DC potential proportional to the AC voltage. Accordingly,by adjusting the applied AC voltage, the capacitance of the tunableelement 3802 can be tuned.

In some embodiments, tunable element 3802 can include other elementssuch as MOSFETs having p-n junctions that exhibit a nonlinearcapacitance. Elements such as MOSFETs can be used for automaticregulation of impedance and/or with the use of control signals from acontrol circuit applied to gates of MOSFETs to regulate their impedance.Such techniques can be used for capacitance tuning of the tunableelement 3802 without utilizing a microcontroller or complicated controlscheme because adjusting the magnitude of the AC voltage (andaccordingly the reverse bias DC voltage) across the tunable element 3802can lead to a change in capacitance. For example, an increase in themagnitude of the AC voltage leads to a decrease in the capacitance. Thiscan allow the tunable element 3802 to automatically tune its capacitanceaccording to a desired voltage and/or protective voltage of a load in areceiver-side circuit.

The foregoing devices and techniques can be used for inductance tuningor capacitance tuning in either or both of a source-side circuit and areceiver-side circuit. The operation frequency of power transfer can be85 kHz, for example. In some embodiments, the operation frequency can bewithin 1% (e.g., within 3%, within 5%, within 10%) of 85 kHz. In certainembodiments, the operation frequency can be 85 kHz or more.

In some embodiments, the operation frequency of power transfer can be145 kHz. The operation frequency can be within 1% (e.g., within 3%,within 5%, within 10%) of 145 kHz. In certain embodiments, the operationfrequency can be 145 kHz or more.

Impedance Matching Networks

The tunable inductors disclosed in the foregoing section can be used ina wide variety of impedance matching networks in wireless power transfersystems. FIG. 13 shows an example of one such impedance matching networkin both the source resonator and the receiver resonator. More generally,however, the tunable inductors disclosed herein can be used in manydifferent networks. Additional examples of impedance matching networksare described in U.S. Patent Application Publication No. 2010/0308939,the entire contents of which are incorporated herein by reference.

Impedance Matching Network Adjustment and Optimization

Impedance matching networks can be adjusted, modified, and/or optimizedto account for a variety of different network components (e.g.,capacitors, inductors, resistors, rectifiers, voltage sources), powertransfer specifications, and operating conditions. To determine suitablevalues of various operating parameters and/or to determine the effectsof adjusting various aspects of a wireless power transfer system'simportant system operating characteristics (such as the magneticcoupling coefficient, k, between resonators, operating efficiency,maximum transferrable power, voltage, and current), the systems can besimulated and optimized.

FIG. 22 shows a schematic diagram of two different wireless powertransfer systems. In the system on the left side of FIG. 22, a deviceresonator receiving power wirelessly is displaced vertically from asource resonator transmitting power wirelessly, with a gap between theseresonators of between 140 mm and 170 mm. A vertical gap in this rangecan correspond, for example, to the gap between a source resonator and avehicle-mounted device resonator in a wireless electric car chargingsystem. The right side of FIG. 22 shows a device resonator verticallydisplaced from a source resonator by a gap of between 150 mm and 220 mm,which can correspond, for example, to a source-device gap in a wirelesspower transfer system for an electric truck or other large electricvehicle.

FIGS. 23-30 illustrate an example of impedance matching networksimulation and adjustment and/or optimization. FIG. 23 shows a schematicdiagram of the impedance matching network topology simulated, whichincludes networks on both the source and device sides of the system. Thesource side includes an inductor L_(S) which couples wirelessly, withcoupling constant k, to an inductor L_(D) on the device side to transferpower from the source side to the device side. The source impedancematching network includes capacitors C_(S1), C_(S2), and C_(S3), and aninductor L_(S3). Capacitor C_(S3) and inductor L_(S3) together form atunable reactance X₃.

The device side includes capacitors C_(D1), C_(D2), and C_(D3), and ACload R_(AC). The topology shown in FIG. 23 is only an example. Moregenerally, a wide variety of different topologies can be simulated,adjusted, and/or optimized. For example, in some embodiments, the deviceside can include a DC load and one or more rectifiers configured toconvert AC power to DC power. The device side can also include one ormore inductors coupled to the rectifier(s). Alternative capacitorarrangements can also be used such as a “pi” capacitor network.

In the system shown in FIG. 23, L_(S) and L_(D) were assumed forpurposes of simulation to be 58.5 μH and 33.2 μH, respectively, althoughthese parameters can take any value according to the particular systemconfiguration. The Q values for the source and device resonators wereassumed to be 300 and 250, respectively, and the coupling constant k wasrestricted to values between 0.12 and 0.31. L_(S), L_(D), and the Qvalues were further assumed to be independent of k. As above, ingeneral, the values for Q and k can be selected as desired according toparticular system configurations.

To optimize the system shown in FIG. 23 (and more generally, systemshaving a wide range of topologies), the impedance matching networks canbe adjusted to maximize an overall figure of merit (FOM). The FOM can bedefined in a variety of ways. For the system shown in FIG. 23, the FOMwas defined as:

FOM=η×(η>0.91?)×(340 V≤V _(bus)≤430 V?)×(5°≤φ≤45° ?)

where η is the efficiency of the system, V_(bus) is the system's busvoltage, and φ is the phase of the complex input impedance to theamplifier in the source.

In the equation for FOM above, the three terms in brackets cancorrespond to Boolean tests which have values of 1 if true, 0 if false.Thus, for example, if η≤0.91, V_(bus) is outside the range from 340 V to430 V, or φ is outside the range from 5° to 45°, the FOM is zero. TheseBoolean conditions effectively restrict the range of values of η,V_(bus), and φ over which the system is simulated by setting FOM=0everywhere else. When the values of these parameters falls within theranges above, FOM=η, with FOM=1 being the maximum FOM value.

In some embodiments, the three terms in brackets in the equation for FOMcan correspond to tests with a smoother (e.g., continuous, rather thandiscrete Boolean) transition between limiting values of 1 and 0, as thetested parameter value goes from satisfying the test to not satisfyingthe test, rather than a strict Boolean test. For example, as V_(bus)goes from more than 340 V to less than 340 V, the value of the testfunction in the FOM may go smoothly from 1 to 0 over a desired voltagerange. It should be appreciated that the range of values of η, V_(bus),and φ can generally be selected as desired to optimize systems ofvarious configurations.

To optimize the system, the FOM is evaluated at each point in k-V_(load)space, where V_(load) represents the range of voltages that the systemprovides to a load connected on the device side. An overall FOM is thencalculated as the integral over the domain in k-V_(load) space definedby 0.125≤k≤0.325 and 300 V≤V_(load)≤400 V, with each point within thedomain having equal weight. In the plots shown in FIGS. 24-30, thisdomain is enclosed by a dashed box.

A variety of different methods for adjustment and optimization of theimpedance matching network shown in FIG. 23 were investigated. Each ofthe plots in FIGS. 24A-F shows FOM calculated for a different type ofimpedance adjustment. The plot in FIG. 24A corresponds to a fixedimpedance matching network with no tuning. The plots in FIGS. 24B and24C correspond to a fixed impedance matching network with frequencytuning in a range from 80-90 kHz, and in a range from 82.5-87.5 kHz,respectively.

The plot in FIG. 24D corresponds to an impedance matching network whereX₃ is continuously tunable on the source side. The plot in FIG. 24Ecorresponds to an impedance matching network where capacitor C₂ istunable on the source side between two discrete values. The plot in FIG.24F corresponds to a fixed impedance matching network with a DC-DCconverter on the device side. The DC-DC converter reduces the variationin V_(load) to zero, so that in FIG. 24F, only k varies. As is evidentfrom the plots in FIGS. 24A-F, frequency tuning with a fixed impedancematching network yields high FOM values (FIGS. 24B and 24C) over a largedomain in k-V_(load) space.

FIGS. 25A-E are plots showing optimal values of the tuning parameter forthe various impedance matching network tuning methods described inconnection with FIGS. 24A-F. FIG. 25A shows optimal values of the tuningfrequency for frequency tuning in the 80-90 kHz band, FIG. 25B showsoptimal values of the tuning frequency for frequency tuning in the82.5-87.5 kHz band, FIG. 25C shows optimal values of the change inreactance for continuous X₃ tuning on the source side, FIG. 25D showsoptimal values of C₂ capacitance for discrete C₂ capacitance tuningbetween two values on the source side, and FIG. 25E shows optimal valuesof V_(load) for a fixed impedance matching network with a DC-DCconverter on the device side.

FIGS. 26A-F are plots showing values of the bus voltage V_(bus) for theabove-described impedance matching network tuning methods. Inparticular, FIG. 26A shows V_(bus) values for a fixed impedance matchingnetwork with no tuning, FIG. 26B shows V_(bus) values for a fixedimpedance matching network with frequency tuning in the 80-90 kHz band,FIG. 26C shows V_(bus) values for a fixed impedance matching networkwith frequency tuning in the 82.5-87.5 kHz band, FIG. 26D shows V_(bus)values for source-side continuous X₃ tuning, FIG. 26E shows V_(bus)values for source-side discrete C₂ tuning between two discrete values,and FIG. 26F shows V_(bus) values for a fixed impedance matching networkwith a DC-DC converter on the device side. As shown in FIGS. 26B and26C, network optimization by frequency tuning yields consistent V_(bus)values over a wide range of the k-V_(load) domain.

FIGS. 27A-F are plots showing values of the input phase φ for theabove-described impedance matching network tuning methods. Inparticular, FIG. 27A shows φ values for a fixed impedance matchingnetwork with no tuning, FIG. 26B shows φ values for a fixed impedancematching network with frequency tuning in the 80-90 kHz band, FIG. 26Cshows φ values for a fixed impedance matching network with frequencytuning in the 82.5-87.5 kHz band, FIG. 26D shows φ values forsource-side continuous X₃ tuning, FIG. 26E shows φ values forsource-side discrete C₂ tuning between two discrete values, and FIG. 26Fshows φ values for a fixed impedance matching network with a DC-DCconverter on the device side.

FIGS. 28A-F are plots showing values of (combined) coil-to-coiltransmission and impedance matching network efficiency for theabove-described impedance matching network tuning methods. Theefficiency values calculated and shown in FIGS. 28A-F do not account forefficiency losses in other system components such as inverters andrectifiers. Specifically, FIG. 28A shows efficiency values for a fixedimpedance matching network with no tuning, FIG. 28B shows efficiencyvalues for a fixed impedance matching network with frequency tuning inthe 80-90 kHz band, FIG. 28C shows efficiency values for a fixedimpedance matching network with frequency tuning in the 82.5-87.5 kHzband, FIG. 28D shows efficiency values for source-side continuous X₃tuning, FIG. 28E shows efficiency values for source-side discrete C₂tuning between two discrete values, and FIG. 28F shows efficiency valuesfor a fixed impedance matching network with a DC-DC converter on thedevice side. As shown in FIGS. 28B and 28C, relatively high efficiencyvalues can be achieved by frequency tuning over a wide range of thek-V_(load) domain.

FIGS. 29A-F are plots showing values of power dissipated in the sourcefor the above-described impedance matching network tuning methods.Specifically, FIG. 29A shows power dissipated for a fixed impedancematching network with no tuning, FIG. 29B shows power dissipated for afixed impedance matching network with frequency tuning in the 80-90 kHzband, FIG. 29C shows power dissipated for a fixed impedance matchingnetwork with frequency tuning in the 82.5-87.5 kHz band, FIG. 29D showspower dissipated for source-side continuous X₃ tuning, FIG. 29E showspower dissipated for source-side discrete C₂ tuning between two discretevalues, and FIG. 29F shows power dissipated for a fixed impedancematching network with a DC-DC converter on the device side. As shown inFIGS. 29B and 29C, power dissipation in the source can be minimized overa wide range of the k-V_(load) domain by frequency tuning.

FIGS. 30A-F are plots showing values of power dissipated in the devicefor the above-described impedance matching network tuning methods.Specifically, FIG. 30A shows power dissipated for a fixed impedancematching network with no tuning, FIG. 30B shows power dissipated for afixed impedance matching network with frequency tuning in the 80-90 kHzband, FIG. 30C shows power dissipated for a fixed impedance matchingnetwork with frequency tuning in the 82.5-87.5 kHz band, FIG. 30D showspower dissipated for source-side continuous X₃ tuning, FIG. 30E showspower dissipated for source-side discrete C₂ tuning between two discretevalues, and FIG. 30F shows power dissipated for a fixed impedancematching network with a DC-DC converter on the device side.

Frequency Tuning in Wireless Energy Transfer Systems

A variety of challenges exist when providing wireless power for chargingelectric vehicles. FIG. 31 is a schematic diagram showing an electricvehicle 3103 with a wireless power receiver 3102 mounted to theunderside of the vehicle (e.g., mounted to the vehicle chassis). Awireless power transmitter 3101 is positioned in the ground or in thefloor of a structure (e.g., in a driveway or a garage) so that whenvehicle 3103 is parked, power transmitter 3101 and power receiver 3102are aligned and power can be transferred from power transmitter 3101 topower receiver 3102. Power receiver 3102 is typically connected to oneor more of the vehicle's batteries and electrical systems, andwirelessly transferred power can be used to charge the vehicle'sbatteries and/or to power the vehicle's electrical systems.

For reference purposes, in the following discussion, the coordinatesystem shown in FIG. 31 is adopted unless otherwise stated. In thiscoordinate system, wireless power transmitter 3101 and wireless powerreceiver 3102 are separated along the z-direction when vehicle 3103 isparked over transmitter 3101. Vehicle 3103 can also be displaced frompower transmitter 3101 along either of the x- and y-directions as well.That is, vehicle 3103 can be displaced relative to power transmitter3101 in the plane of the ground or floor in which power transmitter 3101is positioned. In embodiments, displacements of vehicle 3103 in any ofthe coordinate directions relative to power transmitter 3101 affects theefficiency with which power can be wirelessly transferred between powertransmitter 3101 and power receiver 3102.

In particular, when a vehicle is parked repeatedly over powertransmitter 3101, considerable variation can exist in the position ofthe vehicle relative to the power transmitter. As the position of thevehicle relative to power transmitter 3101 changes in the x-, y-, andz-directions, the range of output power and output voltages from powerreceiver 3102 changes. If vehicle 3103 is displaced sufficiently far inthe x- and/or y-directions (e.g., so that power receiver 3102 is notvertically above (e.g., in the z-direction) power transmitter 101), therange of output power and output voltages from the power receiver 3102can be limited significantly. This limits the ability of a wirelesspower transfer system to efficiently provide charging power to thevehicles batteries, which may require a range of voltages over acharging cycle. As an example, for a power transmitter and receiver thatare aligned in the x-y plane and displaced by 12.5 cm in thez-direction, the wireless power receiver can provide full power acrossan output DC voltage range of 300-360 V. If the power transmitter andpower receiver are displaced from one another by 6 cm in they-direction, the wireless power receiver's DC voltage range is reducedto 300-315 V. To compensate for the reduced voltage range, DC-DCconverters can be used to expand the voltage output range from therectifier in power receiver 3102. However, the use of such converterscan significantly reduce the overall efficiency of power receiver 3102.

FIG. 32 is a schematic view of wireless power transmitter 3101 andwireless power receiver 3102. Wireless power transmitter 3101 typicallyincludes one or more coils 3202 coupled to a power source 3206. As shownin FIG. 32, in some embodiments, coils 3202 can be coupled to powersource 3206 through an optional impedance matching network 3204. Acontroller (e.g., which typically includes one or more electronicprocessors) is coupled to power source 3206 and impedance matchingnetwork 3204.

Wireless power receiver 3102 includes one or more coils 3210 coupled,through optional impedance matching network 3212, to a load 3214. Load3214 can be one or more vehicle batteries, a vehicle electrical systemor circuit, or any other vehicle component that draws electrical power.

Coil(s) 3202 form part of a power transmitting resonator in wirelesspower transmitter 3101, and coil(s) 3210 form part of a power receivingresonator in wireless power receiver 3102. During operation, powersource 3206 delivers power to coil(s) 3202, which generate magneticfields at a frequency determined by power source 1106 under the controlof controller 3208. The frequency is typically chosen to be resonantwith a frequency of the power receiving resonator in wireless powerreceiver 3102, so that the magnetic fields induce a current in coil(s)3210, thereby delivering electrical power to load 3214.

As power transmitter 3101 and power receiver 3102 are displaced from oneanother in the x-, y-, and z-directions, the coupling value k betweenthe power transmitting and power receiving resonators is reduced, andtherefore the reflected impedance of the power receiving resonatorconnected to load 3214 that is seen by power source 3206 changes.Additionally, the load 3214 may also change during operation, forexample as the battery voltage changes during the charge cycle, whichmay change the reflected impedance that is seen by power source 3206.Because the bus voltage within the amplifier of power source 3206 islimited to a maximum value and the current that can be supplied by thepower source is limited to a maximum value, changes in the reflectedimpedance typically limit the range of the power and voltage that can bedelivered to load 3214. For large displacements in any of the coordinatedirections, certain voltages may not be achievable at load 3214, evenwhen power source 3206 is operating at capacity.

In some vehicle charging applications, the frequency of power transferis nominally fixed, e.g., at a value of 85 kHz. Even in fixed frequencysystems, however, the reflected impedance is frequency dependent, and bychanging the frequency at which power is delivered, the reflectedimpedance of power receiving resonator with load 3214 as seen by powersource 3106 can be adjusted. Thus, by adjusting or “tuning” thefrequency of power transfer between power transmitter 3101 and powerreceiver 3102, controller 3208 can compensate for changes in couplingthat occur due to relative displacements between power transmitter 3101and power receiver 3102 in any of the coordinate directions and forchanges in load 3214, allowing higher output voltages at load 3214 to beachieved. As mentioned briefly above, maintaining a relatively widerange of output voltages at load 3214 can be important for loads such asvehicle batteries, which typically require a range of voltages over acharging cycle.

Tuning the frequency at which power is transferred can thus provide anumber of significant operating advantages. For example, adjusting thefrequency by as little as ±5% or less can, in some wireless powertransfer systems, increase the voltage range at load 3214 by as much as100 V. In addition, impedances as seen by the power source 3206 thatwould otherwise be outside the operating range of power transmitter 3101can be changed so that they are within the operating range of thetransmitter. Power transfer at displacements between power transmitter3101 and power receiver 3102 that would otherwise have been too largefor effective power transfer to load 3214 can be achieved, and existingvoltage ranges at previously achievable displacements can be increased.

In some embodiments, increasing the available voltage range at load 3214can also increase the overall system efficiency. Certain wireless powertransfer systems are more efficient when operating at higher voltages.Even when such systems include a DC-DC converter, higher voltage rangescan be made accessible by frequency tuning, thereby improving systemefficiency.

FIG. 33 shows a schematic diagram of a simulated wireless power transfersystem that includes a power transmitter 3101 and a power receiver 3102.At a relative displacement of 10 cm in the z-direction and ±15 cm in thex-direction, when transferring 3300 Watts the achievable voltage rangeat a load connected to power receiver 3102 is 340-400 V withoutfrequency tuning, and 260-400 V with frequency tuning. At a relativedisplacement of 10 cm in the z-direction and ±7.5 cm in the y-direction,when transferring 3300 Watts the achievable voltage range at a loadconnected to power receiver 3102 is 340-400 V without frequency tuning,and 260-400 V with frequency tuning.

For a relative displacement of 12.5 cm in the z-direction and a relativedisplacement of ±12.5 cm in the x-direction, when transferring 3300Watts the voltage range without frequency tuning is 300-315 V, and thevoltage range with frequency tuning is 260-400 V. When the displacementin the x-direction is smaller (e.g., ±10 cm or less), when transferring3300 Watts the voltage range without frequency tuning is 300-360 V, andthe voltage range with frequency tuning is 260-400 V.

For a relative displacement of 12.5 cm in the z-direction and ±6 cm inthe y-direction, when transferring 3300 Watts the voltage range withoutfrequency tuning is 300-360 V, and the voltage range with frequencytuning is 260-400 V. When the displacement in the y-direction is smaller(e.g., ±4.5 cm or less), when transferring 3300 Watts the voltage rangewithout frequency tuning is 300-360 V, and the voltage range withfrequency tuning is 260-400 V.

For even larger relative displacements of 15 cm in the z-direction, novoltage range for displacements of ±12.5 cm or ±10 cm in the x-directionis achievable without frequency tuning, but frequency tuning allowsvoltage ranges of 260-350 V and 260-400 V to be achieved, respectively,when transferring 3300 Watts. For smaller displacements in thex-direction (e.g., ±5 cm or less), when transferring 3300 Watts thevoltage range without frequency tuning is 260-270 V, and the voltagerange with frequency tuning is 260-400 V.

For relatively displacements of 15 cm in the z-direction and ±6 cm or±4.5 cm in the y-direction, no voltage range is achievable withoutfrequency tuning, but voltage ranges of 260-350 V and 260-400 V,respectively, when transferring 3300 Watts can be achieved withfrequency tuning. For smaller displacements in the y-direction (e.g., ±3cm or less), a voltage range of 260-270 V can be attained withoutfrequency tuning when transferring 3300 Watts. The voltage range withfrequency tuning is 260-400 V.

The foregoing measurement results demonstrate that the output voltagerange in wireless power transfer systems can be significantly expandedusing frequency tuning techniques. In some embodiments, these techniqueseven permit wireless power transfer where it would not otherwise occurdue to relatively large displacements between the power transmitter andreceiver. Considered another way, the use of frequency tuning methodspermits a wider range of alignment tolerances between the powertransmitter and receiver in a wireless power transfer system.Flexibility with regard to alignment tolerances can be particularlyimportant in vehicle charging applications, where the alignment betweenthe power transmitter and receiver varies frequently.

FIG. 34 shows a flow chart 3400 that includes a series of example stepsfor implementing frequency tuning in wireless power transfer systems.Not all steps are required in flow chart 400; the example is shown onlyto illustrate a functional implementation of frequency tuning. Inembodiments, a controller (e.g., controller 3208) performs the stepsshown in FIG. 34. In some embodiments, controller 3208 can perform thesteps automatically with no feedback or input from a human operator.

The frequency tuning methods disclosed herein typically use a measuredsignal, which will be referred to subsequently as the “CapDetect”signal, for feedback during frequency adjustment. The CapDetectsignal—which can be used to determine whether the wireless powertransfer system is operating in a capacitive mode, i.e., the impedanceseen by the power source has a negative reactance, —corresponds to themeasured time between the switching edge of the amplifier in powersource 3206 and the zero crossing of the output resonant current fromthe amplifier. In other words, the CapDetect signal corresponds to thephase difference between the voltage and current output signals frompower source 3206. In embodiments, a wireless power transmitter isconfigured for operation at a range of different CapDetect values for aparticular frequency. When the measured CapDetect signal falls outsidethe allowable range of values, controller 3208 can be configured to shutdown the wireless power transmitter (e.g., to prevent damage to internalcomponents from overheating).

FIG. 35 shows a series of measured signals from a wireless powertransmitter. In particular, signal 3502 corresponds to the outputvoltage from the amplifier in power source 3206, signal 3504 correspondsto the output current from the amplifier, and signal 3506 is themeasured CapDetect signal corresponding to the phase difference betweensignals 3502 and 3504.

Returning to FIG. 34, in a first step 3402, the wireless power transfersystem is initialized for frequency tuning. Initialization can include,for example, setting power transmitter to operate at a minimum frequency(e.g., a minimum frequency of 81.38 kHz for 85 kHz systems), setting theamplifier power control phase to a minimum value (e.g., reducing to aminimum power output from power source 32), and/or setting power source3206 to operate at a minimum internal bus voltage.

Next, in step 3404, the shutdown check procedure—which is based on themeasured value of the CapDetect parameter—is disabled, if it exists.Then, in step 3406, the power control phase in power source 3206 ischecked to determine whether it is a maximum value. If the power controlphase is not at a maximum value, the power control phase is increased atstep 3408, and the power control phase is checked again in step 3406 todetermine whether it is now at a maximum value. If instead the powercontrol phase is already maximized, the CapDetect-based shutdown checkprocedure is re-enabled at step 3410.

Then, at step 3412, the power output is measured to determine whetherthe power transmitter is delivering a target amount of power to a load(e.g., load 3214 connected to a wireless power receiver 3102). If thetarget output power level has been reached, control passes to step 3414,and controller 3208 regulates power source 3206 at the target powerlevel. If the target output power level has not been reached, then instep 3416, controller 3208 determines whether the output power isgreater than a pre-determined minimum power level P_(min) (which can be,for example, 500 W). If the output power level is less than P_(min),controller 3208 increases the frequency of the oscillating magneticfield generated by power transmitter 3101 (e.g., by increasing thefrequency of the current delivered to coil(s) 3202), and control returnsto step 3416, where the output power level is checked again.

After the output power level is larger than P_(min), the frequency ofthe field is optimized in step 3420. This optimization step will bediscussed in greater detail subsequently.

Next, the output power at the optimized frequency is measured in step3422 and compared to the target output power. If the output power levelmatches the target power level, control passes to step 3414 and theoutput power is regulated by controller 3208 at the target level.Otherwise, controller 3208 increases the bus voltage in power source3206 in step 3424, and the output power level is again measured andcompared to the target power level in step 3426. If the target powerlevel has been reached, control passes to step 3414.

If the target power level has not been reached, controller 3208 checksthe bus voltage in step 3428 to determine whether it is greater than apre-determined threshold bus voltage V_(th) (e.g., in some embodiments,V_(th) can be 400 V). If the bus voltage remains less than V_(th), thenthe bus voltage is again increased when control passes to step 3424. Ifinstead the bus voltage has reached V_(th), then another frequencyoptimization is performed at step 3430.

Following step 3430, the output power is again measured and compared tothe target output power level in step 3432. If the target power levelhas been reached, control passes to step 3414. Otherwise, controller3208 increases the bus voltage at step 3434, and control returns to step3432.

FIG. 36 is a flow chart 3600 showing a series of steps for regulatingthe output power at a pre-determined power level (e.g., within ±100 W ofa target output power level). The procedure begins in step 3602 andcontroller 3208 checks the output power level in step 3604 to determinewhether it is higher than a pre-determined maximum power level. If it ishigher, controller 3208 checks the bus voltage in step 3606 to determinewhether it is at a minimum value. If it is larger than the minimumvalue, the bus voltage is reduced in step 3608 and control returns tostep 3604.

Alternatively, if the bus voltage is at the minimum value, thencontroller 3208 checks the frequency to determine whether it has reacheda minimum value. If not, the frequency is reduced in step 3612 andcontrol returns to step 3604. If the frequency has instead reached aminimum pre-determined value, then controller 3208 checks to determinewhether the power control phase has reached a minimum value in step3614. If the power control phase has not reached the minimum value, thenthe power control phase is reduced in step 3616. Control then returns tostep 3604 from either step 3614 or step 616.

If the output power has not exceeded the maximum value in step 3604,then the output power level is checked to determine whether it hasexceeded a pre-determined minimum value in step 3618. If it has not,then control returns to step 3604. But if the output power has fallenbelow the minimum value, then a frequency optimization step occurs instep 3620. Next, in step 3622, the output power at the optimizedfrequency is checked against the pre-determined minimum value. If thepower is no longer lower than the minimum value, control returns to step3604. However, if the output power remains low, then control passes tostep 3624 and controller 3208 checks to determine whether the maximumpower control phase value has been reached.

If the power control phase is not yet at a maximum value, the powercontrol phase is reduced in step 3626, and control returns to step 3622.Alternatively, if the maximum power control phase value has beenreached, then controller 3208 checks to determine whether apre-determined maximum bus voltage has been reached in step 3628. If thebus voltage is less than the maximum bus voltage, the bus voltage isincreased at step 3630 and control returns to step 3622. If the maximumbus voltage has been reached in step 3628, control returns to step 3604.

Flow charts 3400 and 3600 both include frequency optimization steps.FIG. 37 shows a flow chart 3700 that includes a series of steps foroptimizing the frequency at which magnetic fields are generated by powertransmitter 3101 (and therefore, the frequency at which power isdelivered to load 3214 connected to power receiver 3102).

In embodiments, the procedure shown in flow chart 3700 optimizes thefrequency by slowing increasing the frequency and measuring thecorresponding value of the CapDetect parameter, until the point wherethe measured value of CapDetect passes through a clear minimum value. Atthat point, the procedure terminates, as the frequency setting thatgenerates a minimum CapDetect value has been determined.

In step 7302, controller 3208 ensures that the frequency is betweenpre-determined minimum and maximum values by adjusting the frequency tobe equal to the minimum value if it is currently smaller, and adjustingthe frequency to be equal to the maximum value if it is currentlylarger. Next, in step 3704, controller 3208 stores the current CapDetectvalue, sets a flag (hereinafter the “CapDetect_decrease” flag) to avalue of false to indicate that the CapDetect value has not decreased,and sets a counter (hereinafter the “HighCap” counter)—which counts thenumber of times that the value of CapDetect has exceeded apre-determined high value limit—to a value of zero.

Next, in step 3706, controller 3208 increases the frequency if it isless than the maximum pre-determined value. Then the value of CapDetectis measured in step 3708, and in step 3710, controller 3208 checks todetermine whether the pre-determined high frequency limit has beenreached, which includes determining whether the current value ofCapDetect is less than or equal to the immediate previously measuredvalue, and whether the current frequency has reached the maximumfrequency value, and whether the HighCap counter has exceeded a limitingvalue (e.g., 3). If true, the procedure terminates at step 3712.

Otherwise, in the next step 3714, controller 3208 performs a “CapDetect1” check that includes determining whether the current value ofCapDetect is larger than the immediate previously measured value, andthe value of the CapDetect_decrease flag is true, and whether theHighCap counter has exceeded the limiting value. If true, the frequencyis reduced at step 3716 and the procedure terminates.

The procedure otherwise continues at step 3718, where controller 3208performs a “CapDetect 2” check that includes determining whether thecurrent value of CapDetect is larger than the immediate previouslymeasured value, and whether the HighCap counter is less than or equal tothe limiting value. If true, HighCap counter is increased at step 3720,and control returns to step 3708.

If the CapDetect 2 check returns a false value, controller 3208 performsa “CapDetect 3” check at step 3722, which includes determining whetherthe current CapDetect value is less than or equal to the immediatepreviously measured value, and the current frequency is less than thepre-determined maximum frequency value. If true, controller 3208increases the frequency at step 3724 and, at step 3726, sets HighCapcounter to zero, sets the CapDetect_decrease flag to true, and storesthe current CapDetect value. Control then returns to step 3708.

If the CapDetect 3 check returns a false value, then controller 3208performs a “CapDetect 4” check at step 3728, which includes determiningwhether the current CapDetect value is larger than the immediatepreviously measured value, and whether the measured CapDetect hasdecreased (i.e., whether CapDetect_decrease is false), and whether theHighCap counter is larger than the limiting value. If true, thefrequency is reduced at step 3730, and at step 3732, the value ofCapDetect is stored and HighCap counter is set to zero.

Control then passes to step 3734, where controller 3208 again measuresthe CapDetect value. In step 3736, controller 3208 performs a “CapDetect5” check, which includes determining whether the current CapDetect valueis less than or equal to the immediate previously measured value, andwhether the current frequency is less than the pre-determined minimumfrequency, and the HighCap counter has exceeded the limiting value. Iftrue, the procedure terminates.

If false, controller 3208 performs a “CapDetect 6” check at step 3738,which includes determining whether the current CapDetect value isgreater than the immediate previously measured value, and whether theHighCap counter is greater than the limiting value. If true, thefrequency is increased at step 3740, and the procedure terminates.

Otherwise, controller 3208 performs a “CapDetect 7” check at step 3742,which includes determining whether the current CapDetect value isgreater than the immediate previously measured value, and whether theHighCap counter is less than or equal to the limiting value. If true,the HighCap counter is incremented in step 3744, and control returns tostep 3734.

If false, controller 3208 performs a “CapDetect 8” check at step 3746,which includes determining whether the current CapDetect value is lessthan or equal to the immediate previously measured value, and whetherthe current frequency is greater than the pre-determined maximumfrequency value. If false, control returns to step 3734.

If true, the frequency is decreased at step 3748, and then at step 3750,controller 3208 stores the CapDetect value and sets the HighCap counterto zero. Control then returns to step 3734.

Thus, by measuring the value of CapDetect (that is, the phase differencebetween the output voltage and current signals produced by power source3206), the frequency at which power is delivered can be tuned. Asdiscussed above, tuning the frequency allows relative displacementsbetween power transmitter 3101 and power receiver 3102 to becompensated, which is particularly useful in vehicle chargingapplications.

In addition, frequency tuning allows for a significant increase in theoutput voltage range delivered to a load connected to power receiver3102. Operating with a wider output voltage range can increase theefficiency of the wireless power transfer system. Further, operatingwith a wider output voltage range can permit delivery of power tocertain types of loads—such as vehicle batteries—that typically demand arange of voltages during power consumption or during a charging cycle.

Frequency Tuning and Figure of Merit

In wireless power transfer systems, nonlinear optimizations can beperformed to select certain system components to maximize a figure ofmerit (FOM) defined for the system. The FOM is a numerical value that iszero or small if certain system requirements are not satisfied (e.g.,voltages or phases that are not within specified ranges). Otherwise, theFOM may represent a measure of efficiency of the system. Other systemparameters that may be important for particular systems, such ascurrents or voltages on components such as capacitors, inductors,diodes, electronic switches, etc., may also be included in thedefinition of FOM. As discussed previously, in general, the FOM isevaluated and optimized in a two-dimensional k-V_(load) space (that is,a space defined by a coordinate system in the magnetic couplingcoefficient, k, between source and receiver resonators, and the voltageat the load coupled to the receiver resonator). Various system variablescan change in k-V_(load) space. For example, resonator inductances canchange as a function of k. These changes can be incorporated into theFOM optimization because the FOM is evaluated at each point in thespace.

To evaluate FOM, impedance matching network topologies are selected andvarious system parameters and constraints are defined. For example, aC1/C2/C3/L3 matching network can be used in the power transmissionapparatus (the “Source” network) and a C2/C3 matching network can beused in the power receiving apparatus (the “Device” network), as shownin FIG. 38. The parameter V_(load) can be constrained within a range of,for example, 290-353 V, and the bus voltage V_(bus) in the powertransmission apparatus can be constrained to a range of, for example,390-440 V. The frequency of the transmitted power can be constrainedwithin a range from 81.88-89.5 kHz, for example. With constraints inplace, the FOM can be evaluated as

FOM=η×(η>0 0.90?)×(V _(bus,min) ≤V _(bux) ≤V _(bux,max)?)×(5°≤ϕ≤45° ?)

where, as described previously, η is the efficiency of the system,V_(bus) is the system's bus voltage, and φ is the phase of the complexinput impedance to the amplifier in the source. Note that the aboveequation for FOM differs slightly from the FOM equation discussedpreviously, in that a different test value for η is used in the firstbracketed term, and in the second bracketed term, the generalized limitsV_(bus,min) and V_(bus,max) are used in place of specific values forthese parameters. In general, the specific values for the variousparameters in the bracketed test terms can be selected as desired tosimulate, adjust and optimize wireless power transfer systems having awide variety of configurations.

Also, as discussed above, FOM optimization techniques can be combinedwith the frequency tuning methods disclosed herein to ensure thatwireless power transfer systems operate at high efficiency. To optimizea system, for example, FOM calculations can be performed iteratively. Ateach iteration in the optimization process, impedance matching networksare selected, and then for each point in k-V_(load) space, an operatingfrequency is selected. At each point in k-V_(load) space, FOM is thencalculated, and individual FOMs can be averaged together (or combined inother ways, such as by computing the median and/or applying otherconstraints).

The method by which the operating frequency is selected can affect theresults of the FOM-based optimization. In some embodiments, thefrequency that is selected is the one that maximizes the calculated FOM.However, determining that frequency value can be difficult in practice.Thus, in certain embodiments, the frequency that is selected for the FOMoptimization is the frequency that minimizes the phase in power source3206 (i.e., the frequency that results in the smallest CapDetect value,as discussed in connection with FIG. 37). Frequency values selected inthis manner typically yield FOM values that are nearly as large as thosederived from pure optimization based on the frequency.

Hardware and Software Implementation

The steps described above in connection with various methods forinductance tuning, signal generation and detection, and logicoperations, can be implemented in electrical circuits, in logic units,and/or in one or more electronic processors executing computer programsgenerated using standard programming techniques. Such programs aredesigned to execute on programmable computers, processors, orspecifically designed integrated circuits, each comprising an electronicprocessor, a data storage system (including memory and/or storageelements), at least one input device, and least one output device, suchas a display or printer. The program code is applied to input data(e.g., measured and/or generated signals) to perform the functionsdescribed herein and generate output information (e.g., controlsignals), which is applied to one or more circuit components such astunable inductors, signal generators, and detectors. Each such computerprogram can be implemented in a high-level procedural or object-orientedprogramming language, or an assembly or machine language. Furthermore,the language can be a compiled or interpreted language. Each suchcomputer program can be stored on a computer readable storage medium(e.g., CD ROM or magnetic diskette) that when read by a computer cancause the processor in the computer to perform the analysis and controlfunctions described herein. Electronic processors can, in general, beconfigured through software instructions to perform any of the methodsteps, analysis functions, and control functions disclosed herein.

Other Embodiments

Although the foregoing disclosure focuses largely on attributes andfeatures of the source resonator in a wireless power transfer system,the features, steps, and systems, and devices disclosed herein are alsogenerally applicable to the receiver resonator in a wireless powertransfer system. For example, the receiver resonator can include one ormore impedance matching networks that can include any of the tunableinductors disclosed herein.

Additional features and examples of wireless power transfer systems thatimplement impedance tuning are disclosed, for example, in U.S. PatentApplication Publication No. 2011/0193416, the entire contents of whichare incorporated herein by reference.

A number of embodiments have been described. Nevertheless, it will beunderstood that various modifications may be made without departing fromthe spirit and scope of the disclosure. Accordingly, other embodimentsare within the scope of the following claims.

What is claimed is:
 1. A wireless power transfer system, comprising: apower transmitting apparatus configured to wirelessly transmit power; apower receiving apparatus connected to an electrical load and configuredto receive power from the power transmitting apparatus; and a controllerconnected to the power transmitting apparatus and configured to: receiveinformation about a phase difference between output voltage and currentwaveforms in a power source of the power transmitting apparatus; andadjust a frequency of the transmitted power based on the measured phasedifference.